Power conversion device for electric motor excitation


H02P27/08 - CONTROL OR REGULATION OF ELECTRIC MOTORS, GENERATORS, OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS (structure of the starter, brake, or other control devices, see the relevant subclasses, e.g. mechanical brake F16D, mechanical speed regulator G05D, variable resistor H01C, starter switch H01H; systems for regulating electric or magnetic variables using transformers, reactors or choke coils G05F; arrangements structurally associated with motors, generators, dynamo-electric converters, transformers, reactors or choke coils, see the relevant subclasses, e.g. H01F, H02K; connection or control of one generator, transformer, reactor, choke coil, or dynamo-electric converter with regard to conjoint operation with similar or other source of supply H02J; control or regulation of static converters H02M)
H02P27/06 - CONTROL OR REGULATION OF ELECTRIC MOTORS, GENERATORS, OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS (structure of the starter, brake, or other control devices, see the relevant subclasses, e.g. mechanical brake F16D, mechanical speed regulator G05D, variable resistor H01C, starter switch H01H; systems for regulating electric or magnetic variables using transformers, reactors or choke coils G05F; arrangements structurally associated with motors, generators, dynamo-electric converters, transformers, reactors or choke coils, see the relevant subclasses, e.g. H01F, H02K; connection or control of one generator, transformer, reactor, choke coil, or dynamo-electric converter with regard to conjoint operation with similar or other source of supply H02J; control or regulation of static converters H02M)

FIELD: electricity.

SUBSTANCE: in the power conversion device the second control unit (100) includes the current control command shaping unit (10) shaping the electric motor (6) current control command based on the torque control command T*, the voltage amplitude index calculation unit (150) that calculates the voltage amplitude index (PMF-modulation factor) based on the current control command, the current control commands adjustment unit (80) shaping the value of current control commands adjustment dV based on the PMF-modulation factor and the electric motor (6) frequency FINV and the unit (50) for shaping pulse duration modulation signals/voltage control commands including the ripple suppression signal shaping unit shaping the ripple suppression signal based on DC voltage EFC to suppress the ripple component of the power source 2f-component for shaping a gate signal ( pulse duration modulation signal) into the inverter.

EFFECT: ensuring suppression of the power source 2f-component combined with simultaneous suppression of current overload shaping or excessive torque ripple in an AC electric motor wherein a single-pulse mode is used.

20 cl, 14 dwg

 

The technical field

The present invention relates to a device for converting power for excitation of the motor, suitable for motor control AC.

Prior art

In recent years, the AC motor is used as the power source in the fields of technology, production machines and appliances, as well as in the areas of road transport, vehicles, motor, etc. to bring the AC motor, requires a source of DC power or AC power supply. In General, the device power conversion for the excitation of the motor, to which a source of DC power is used as the input power source has a configuration in which the device power conversion accepts power input of DC voltage originating from the source of DC power, generates an AC voltage having an arbitrary frequency by the inverter circuit and energizes the AC motor. In General, the device power conversion, for which the source of AC power is used as the input power source has a configuration in which the device is of preobrazovaniya power includes a Converter circuit on the input side, once converts the AC voltage received by the Converter circuit, the DC voltage and supplies this DC voltage in the inverter circuit to energize the AC motor.

The configuration etc. of the NAT device and power for excitation of the motor is illustrated in the design of the power conversion for the excitation of the motor used for electric Railways with alternating current, as an example. The tension in the wires as a source of AC power is single-phase alternating voltage of 20-25 kV. This is a single-phase AC voltage is reduced to approximately 1-2 kV by a transformer and then is introduced into the circuit of the Converter device conversion power for excitation of the motor. Converter circuit accepts input power single-phase voltage in 1-2 kV AC, converts single-phase AC voltage into a DC voltage approximately 1500-3000 In and outputs the DC voltage in the inverter circuit.

It is known that the DC voltage as the output for the Converter circuit includes a ripple frequency component at twice the frequency of the source p is Tania alternating current (hereinafter referred to as "2f-component of the power source"). When the frequency of the AC motor is located near this 2f-component of the power source, it is likely that the electric current of the AC motor is changed in the direction of overcurrent, or large ripple occurs in the torque of the AC motor, which prevents safe operation.

Patent document 1 discloses that such 2f-component of the power source included in the DC voltage, is extracted, and the pulse width of pulse-width modulation circuit of the inverter is adjusted to suppress the influence 2f-component of the power source.

Patent document 1: Japanese laid patent application No. S56-49693

A brief statement of the substance of the invention

However, the control for suppressing 2f-component of the power source disclosed in patent document 1 cannot be applied to all examples of applications. For example, to maximize the applied voltage to the AC motor, it is difficult to apply control to the vehicle with the electric motor and the like, which selects and uses the so-called single-pulse mode as the condition of the switching circuit of the inverter.

Single-pulse mode is for use of the switching status in which the number of pulses of the s, included in the half-period of the output line voltage of the inverter is equal to one. However, in the work area in single-pulse mode, you cannot adjust the width of the pulse. If the technology of patent document 1 is applied to a vehicle with an electric motor and the like, which selects and uses single-pulse mode, a problem arises in that the AC motor generates a current overload, or there is too much ripple torque. Therefore, it is difficult to apply the technology of patent document 1, the main aspect of which is to regulate the pulse width of pulse-width modulation scheme of the inverter to the vehicle with the electric motor and the like, which selects and uses monopulse mode.

The present invention is to provide a device of converting power for excitation of the motor, which provides control suppression for 2f-component of the power source while suppressing the formation of overcurrent or excessive ripple torque in the AC motor in the example application, in which single-pulse mode is selected and used as the switching condition of the inverter circuit is.

In order to solve the above problems and achieve the above objectives, the device power conversion for the excitation of the electric motor according to one aspect of the present invention is designed in such a way that includes: a first conversion unit capacity, which is connected to a source of AC power and converts the AC voltage from the power source AC into a DC voltage; a second conversion unit capacity, which is connected to the first conversion unit power and converts the DC voltage into AC voltage, and outputs the AC voltage to the AC motor; a first control unit that controls the first power conversion power; and a second control unit that controls the second power conversion power, the second control unit includes: a forming unit command current, which generates on the basis of at least the control command torque command current control for AC motor; a unit for computing the indices of the voltage amplitude, which calculates on the basis of control commands by the current index of the amplitude of the voltage that must be applied to the AC motor; block R is the regulation of the control current, which generates, based on at least the index of the voltage amplitude and frequency of the AC motor, the control value of the control current to the control command of the control current; and a signal pulse suppression, which forms the basis of the DC voltage signal pulse suppression, and the second control unit generates, based on the control signal including the control command current, which is regulated by the magnitude of the control commands of the control current and the signal suppression pulse, the signal pulse-width modulation in the second block power conversion and outputs the signal pulse-width modulation.

Additionally, the device power conversion for the excitation of the electric motor according to another aspect of the present invention is designed in such a way that includes: a first conversion unit capacity, which is connected to a source of AC power and converts the AC voltage from a source of AC power into a DC voltage; a second conversion unit capacity, which is connected to the first conversion unit power and converts the DC voltage into AC voltage and outputs the AC voltage to the electrode is the " alternating current; a first control unit that controls the first power conversion power; and a second control unit that controls the second power conversion power, the second control unit includes: a forming unit command current, which generates on the basis of at least the control command torque command current control for AC motor; and a computing unit indexes voltage amplitude, which calculates on the basis of control commands by the current index of the amplitude of the voltage that must be applied to the AC motor, and the first control unit includes: a forming unit control commands DC current, which generates a command to control the DC voltage, which is the target value of the DC voltage; and a control unit DC voltage, and performs control to ensure coincidence of the constant voltage control by DC voltage with each other, and when the frequency of the AC motor is in a predefined range and the output voltage of the second power conversion power is set as a predetermined value lower than the maximum voltage, that is can be displayed in accordance with the DC voltage, the set of control commands DC voltage generates and outputs a command to control the DC voltage to ensure coincidence of the output voltage of the second power conversion power with a predefined value.

With the help of the device power conversion for the excitation of the motor according to the present invention, the signal pulse-width modulation for the second conversion unit power is generated according to the control signal including the control command current, which is regulated by the magnitude of the control current command to the control command of the current control and signal surge management component of the ripple 2f-component of the power source. Therefore, provided such an advantage that it is possible to manage suppression for 2f-component of the power source while suppressing the formation of overcurrent or excessive ripple torque in the AC motor in the example application, in which single-pulse mode is selected and used as the switching condition of the inverter circuit.

Brief description of drawings

The invention is further explained in the description of the preferred variants of the embodiment is about references to the accompanying drawings, on which:

Figure 1 depicts a diagram of a configuration example of the NAT device and power for excitation of the motor in the first embodiment of the present invention.

Figure 2 depicts a diagram of example of a detailed configuration of the processing unit command current control shown in figure 1.

Figure 3 depicts a diagram of example of a detailed configuration of the signal pulse-width modulation/control voltage, shown in figure 1.

Figure 4 depicts a diagram of example of a detailed configuration of the computing unit signal pulse suppression, shown in figure 3.

Figure 5 depicts a diagram of example, the internal state of the computing unit signals the suppression of ripples in the first embodiment.

6 depicts a diagram of an example of a detailed block configuration of the control commands control the current, shown in figure 1.

Fig.7 depicts the consolidation scheme unit generating control commands modulation depth, is shown in Fig.6.

Fig depicts a chart for explaining the relationship between output frequency FINV of the inverter and the transition factor PMF modulation, the transition pulse mode and the switching operation of the selection switch (see figure 3) in the first embodiment.

Fig.9 depicts a graph of total characteristics control the population synchronous motors with permanent magnets in the first embodiment of the present invention and the example of the prior art.

Figure 10 depicts a diagram to explain the control modes in the first embodiment of the present invention.

11 depicts a diagram of an example configuration of the NAT device and power for excitation of the motor in the second embodiment of the present invention.

Fig depicts a diagram of a first example configuration of the processing unit commands the DC voltage in the second embodiment, shown in 11.

Fig depicts a diagram of a second example configuration of the processing unit commands the DC voltage in the second embodiment, shown in 11.

Fig is a diagram for explanation of operating modes in the example in the prior art.

Description of the preferred embodiments of the invention

Embodiments of the device power conversion for the excitation of the motor according to the present invention are explained in detail below with reference to the accompanying drawings. The present invention is not limited to the implementation, as discussed below.

The first option exercise

Figure 1 is a diagram of a configuration example of the NAT device and power for excitation of the motor in the first embodiment, n the present invention. Figure 1 shows an example of the configuration of the NAT device and power for excitation of the motor, which controls a synchronous motor with permanent magnets as an AC motor.

In figure 1 the device 300 conversion power for excitation of the motor in the first embodiment, includes a Converter 220, which acts as a first conversion unit capacity, which adopts the single-phase AC voltage source 230 AC power and converts single-phase AC voltage into a DC voltage, the capacitor 1, which acts as the source of DC power, an inverter 2, which acts as the second conversion unit power, which converts the DC voltage of the capacitor 1 in the AC voltage having an arbitrary frequency, and the electric motor 6 AC (hereinafter simply referred to as "motor"). As a Converter 220 single-phase two-level PWM inverter, single-phase three-level PWM Converter, etc. is appropriate for the application. As the inverter 2 is suitable for use of the inverter based on the type of voltage, such as three-phase two-level PWM inverter is whether three-phase three-level PWM inverter. Since the configuration of the power circuits of both of the inverter 220 and inverter 2 are well known, detailed explanation of the inverter 220 and the inverter 2 is omitted.

The source 230 AC power is the power source that outputs, for example, single-phase voltage in 1-2 kV AC. The Converter 220 is a block voltage Converter that accepts a single-phase AC voltage as input, converts the single-phase AC voltage, for example, the DC voltage approximately 1500-3000 In and outputs the DC voltage to the condenser 1. The DC voltage (voltage of the capacitor 1) as the output of the inverter 220 includes approximately 5% ripple frequency component at twice the frequency of the power supply source 230 AC power (hereinafter called "2f-component of the power source").

The Converter 220 as the first conversion unit power adopts the single-phase AC voltage source 230 AC power, converts the single-phase AC voltage into a DC voltage and outputs the DC voltage to the condenser 1. As a Converter 220 may be used a so-called PWM Converter, the cat is who performs the conversion from AC to DC using a switching element (not shown), such as IGBT (integrated gate bipolar transistor insulated gate). Since the configuration of the power Converter circuit 220 is well known, a detailed explanation of the inverter 220 is omitted.

The sensor 214 current, which detects the input current from the source 230 AC power is placed in the device 300 conversion power for excitation of the motor. The input current IS detected by a sensor 214 current is introduced into the first control unit 200. The control signal CG to control the switching element of the inverter 220 is formed by the first control unit 200 and output to the inverter 220.

The voltage sensor 8, which detects the voltage (hereinafter referred to as "capacitor voltage") EFC of the capacitor 1, is placed in the device 300 conversion power for excitation of the motor. On output lines connecting the inverter 2 and the motor 6 are the sensors 3, 4 and 5 of the current, which detects electric currents iu, iv and iw flowing to the output line. The rotation sensor 7, which detects the signal (mechanical angle θm of the rotor), representing the state of rotation of the rotor, is placed in the motor 6. These detection signals of the sensors 3, 4 and 5 of the current and the rotation sensor 7 are input to the second unit 100 controls.

Can be used the sensor system of rotation, which calculates the position signal from detektirovanie or estimated voltage value, current value, etc. of the electric motor 6 instead of signal (position signal)received from the rotation sensor 7. In this case, the rotation sensor 7 is optional. In other words, receiving a status signal of the rotation is not limited to the use of the rotation sensor 7.

The sensors 3, 4 and 5 supply shall be installed, at least, only in two phases. In this case, the current in the remaining one phase can be obtained by calculation based upon the output signals of the current sensors in two phases. The output current of the inverter 2 can be reproduced and obtained using DC inverter 2.

The gate signals U, V, W, X, Y, and Z, generated by the second unit 100 controls, introduced in the inverter 2. A switching element included in the inverter 2, is subjected to PWM control. As the inverter 2 PWM voltage inverter is suitable for use. Because the configuration of the inverter 2 is well known, a detailed explanation of the inverter 2 is omitted.

The configuration of the second unit 100 controls explained below. As shown in figure 1, the command T* torque control is entered in the second block 100 management of the not shown external device control. E is from the second block 100 management is a component, having a function to control the inverter 2 so that the generated torque T of the electric motor 6 coincides with the input command T* torque control. The second block 100 includes a block 10 forming teams the current control block 150 calculating indexes amplitude voltage calculation unit 40 of governors of the phase angles, the block 50 a signal pulse-width modulation/control voltage, the block 80 of the control commands control the current block 69 calculate the angular frequency of the inverter unit 95 calculating the reference phase angles and the block 90 conversion of three-phase coordinates on the dq-axes. Block 150 calculating indexes amplitude voltage includes a block 20 of the current control d-axis unit 21 to calculate the junction q-axis unit 22 to calculate the noise immunity d-axis unit 23 of the control current of q-axis and the block 30 calculating the coefficients of the modulation.

Block 95 of calculating the reference phase angles calculates the reference phase angle θe of the mechanical angle θm of the rotor. Block 90 conversion of three-phase coordinates on the dq-axes generates a current id d-axis current iq of the d-axis of the three-phase current iu, iv and iw detected by the sensors 3, 4 and 5 of the current, and the reference phase angle θe. Block 69 calculate the angular frequency of the inverter calculates the output angular frequency ω of the inverter from the reference phase angle θe. Block 10 f is Mirovaya control commands shock generates a command id* current control d-axis and the command iq* of the current control q-axis from the command T* torque control, input from outside, and the regulating values dV command of the current control.

Unit 20 controls the current d-axis machines deflection current id between the command id* current control d-axis current id d-axis proportional-integral regulation and generates an error pde current d-axis. Unit 21 to calculate the junction of the q-axis calculates the forward voltage vqFF q-axis of the commands id* current control d-axis and the output angular frequency ω of the inverter. Block 22 calculation of noise immunity d-axis calculates the forward voltage vdFF d-axis from the command iq* of the current control q-axis and the output angular frequency ω of the inverter. Unit 23 of the control current of q-axis machines deviation diq current between the command ip* current control q-axis current iq of the q-axis proportional-integral regulation and generates an error pqe current q-axis. Block 30 calculation of coefficients calculates the modulation factor PMF modulation of commands vd* control voltage d-axis, which is the sum of the errors of the dpe current d-axis and forward voltage vdFF d-axis command vq* control q-axis voltage, which is the sum of the error pqe current q-axis and forward voltage vqFF q-axis reference phase angle θe and the voltage EFC on the capacitor.

Unit 40 calculation of governors of the phase angles calculates managing the phase angle θ of the commands vd* control d-axis voltage commands vq* control voltage is of the q-axis and the reference phase angle θe. The block 80 of the control commands control the current forms of regulating the value of the dV command current control of the ratio PMF modulation and output frequency FINV of the inverter. The block 50 a signal pulse-width modulation/control voltage generates the coefficient of PMF modulation, the control phase angle θ and the output frequency FINV of the inverter gate signals U, V, W, X, Y, and Z in the inverter 2.

According to the functions of the components, configured as explained above, the block 150 calculating indexes voltage amplitude generates a factor PMF modulation command vd* control d-axis voltage and the command vq* control q-axis voltage using deviations did current, forward voltage vqFF q-axis forward voltage vdFF d-axis deviation diq current, voltage EFC on the capacitor and the reference phase angle θe, displays the ratio PMF modulation in block 50 of the signal pulse-width modulation/control voltage and outputs the control command of the d-axis voltage vd*a and command vq* control q-axis voltage in block 40 the calculation of governors of the phase angles.

According to the functions of the components, configured as explained above, the second block 100 control generates the gate signals U, V, W, X, Y, and Z using a mechanical angle θm of the rotor, three-phase currents iu, iv and iw, team * torque control and voltage EFC on the capacitor, and outputs the gate signals U, V, W, X, Y, and Z in the inverter 2.

Detailed configuration and operation control units, explained above, is explained below. First, the block 95 of calculating the reference phase angles calculates on the basis of the following formula reference phase angle θe as the electrical angle mechanical angle θm of the rotor.

θe=θm*PP(1),

where PP is the number of pairs of poles of the motor 6.

Block 90 conversion of three-phase coordinates on the dq-axes generates, on the basis of the following formula current id d-axis current iq of the q-axis of the three-phase currents iu, iv and iw and the reference phase angle θe.

(2)

Block 69 calculate the angular frequency of the inverter calculates, on the basis of the following formula output angular frequency ω of the inverter by differentiating the reference phase angle θe.

ω=dθe/dt(3)

When the output angular frequency ω of the inverter is calculated, the output frequency FINV of the inverter is obtained by dividing the output of the angular frequency ω of the inverter onalso evaluated.

Detailed configuration and works the block 10 forming teams the current control is illustrated below in relation to figure 2. Figure 2 is a diagram of an example of a detailed configuration of the block 10 forming teams current control shown in figure 1.

Block 10 forming teams the current control is a component having the function of forming, on the basis of the command T* torque control input from the outside, the command id* current control d-axis and command iq* of the current control q-axis. Block 10 forming teams the current control includes a block 11 forming control commands reference current of the d-axis unit 15 forming teams current control q-axis and the adder 14. Examples of the formation method of the command id* current control d-axis and command iq* of the current control q-axis include the method of controlling the maximum torque/current for the formation of maximum torque with a certain electric current and the control method based on the maximum efficiency to maintain the efficiency of the motor at the maximum. These optimal control methods are methods of performing control using the motor rotation speed, the absolute value of the output torque, etc. as parameters so that the actual motor current 6 coincides with a predefined calculation formula or the optimal control command current component of the torque command iq* is the Board's current q-axis) and the command current control component of the magnetic flux (command id* current control d-axis), obtained by storing in a table in advance.

In block 10 forming teams current control according to this variant implementation, as shown in figure 2, the command T* torque control is entered in block 11 forming control commands reference current d-axis, and is formed command id1* control the base current of d-axis as a first control command current d-axis. As the formation method of the command id1* control the base current of d-axis known way to control the maximum torque with which the electric motor 6 can generate the desired torque with minimum current. For example, there is a way to obtain on the basis of the command T* torque control optimal command id1* control the base current of d-axis when accessing the map and the method of obtaining the optimal command id1* control the base current of d-axis according to an arithmetic formula. In both methods unit 11 generate control commands reference current d-axis can be configured using various well-known technologies. Therefore, detailed explanation is omitted.

Command id1* control the base current of d-axis formed through the block 11 forming control commands reference current d-axis, is introduced into the adder 14 and is summed with the regulatory value dV team is the Board's current, whereby is formed command id* current control d-axis as a second management command current d-axis. Regulating the value of the dV command current control mainly takes a negative value and gives the correction in the negative direction for the command id1* control the base current of d-axis. Explaining in more detail, the regulatory value dV command current control acts as a control output signal to perform a so-called flux control field weakening to increase the command id* current control d-axis in the negative direction, the formation of magnetic flux in the direction in which the magnetic flux generated by the permanent magnet included in the electric motor 6, is suppressed, and the weakening of the flux linkage of the motor 6, to lower the voltage of the electric motor 6. Regulating the value of the dV command current control is a control output signal generated by block 80, the control commands of the control current. Detailed configuration of the block 80 of the control commands of the current control is illustrated below.

Command id* current control d-axis is shown in block 150 calculating indexes voltage amplitude as the output signal of the block 10 forming teams current control and, on the other hand, is entered in block 15 f is Mirovaya commands control the current q-axis. In module 15 of the formation of the current control q-axis command iq* of the current control q-axis as the first command current control q-axis is formed from a command id* current control d-axis and command T* torque control. As the formation method of the command iq* of the current control q-axis, similar to the formation method of the command id1* control the base current on d-axis, provides a method for obtaining the optimal command iq* of the current control q-axis when accessing the map and the method of obtaining the optimal command iq* of the current control q-axis according to the calculation formula. In both methods unit 15 forming teams current control q-axis can be configured using various well-known technologies. Therefore, detailed explanation is omitted.

The operation unit 150 calculating indexes voltage amplitude is explained next. Again referring to figure 1, block 23 controls the current q-axis forms on the basis of formula (4) error pqe current q-axis obtained by proportional-integral gain difference between the command iq* of the current control q-axis current iq of the q-axis. Unit 20 controls the current d-axis forms on the basis of formula (5) error pde current d-axis obtained by proportional-integral gain difference between the command id* current control d-axis current id d-axis.

pqe=(K1+K2/s)•(iq*-iq)(4)
pde=(K3+K4/s)•(id*-id)(5)

In the above formulas K1 and K3 represent the proportional gain, and K2 and K4 are an integral gain.

As required, the block 150 calculating indexes voltage amplitude may be a control system that can choose either used or not pqe and pde control (i.e. set or no value pqe and pde equal to zero).

Block 22 calculation of noise immunity d-axis calculates, based on the formula (6) forward voltage vdFF d-axis. Unit 21 to calculate the junction on the q-axis calculates on the basis of formula (7) forward voltage vqFF q-axis.

vdFF=(R1+s•Ld)•id*-ω•Lq•iq*(6)
vqFF=(R1+s•Lq)•iq*+ω•(Ld•id*+ϕa)(7)

In the above formulas, R1 represents the resistance of the primary winding (Ω) of the electric motor 6, Ld is the inductance in d-axis (H), Lq is an inductance in q-axis (H), ϕarepresents the magnetic flux of the permanent magnet (Wb) and s is the operator of differentiation.

Block 30 calculating the coefficients of the modulation calculates on the basis of the following formula coefficient is ecient PMF modulation index amplitude of the voltage commands vd* control voltage d-axis, which is the sum of the error pde current d-axis and forward voltage vdFF d-axis command vq* control q-axis voltage, which is the sum of the error pqe current q-axis and forward voltage vqFF q-axis reference phase angle θe and the voltage EFC on the capacitor.

PMF=VM*/VMmax(8)

VMmax and VM* in the formula (8) are represented by the following formulas:

VMmax=(/π)•EFC(9)
MV*=sqrt(vd*2+vq*2)(10).

Factor PMF modulation indicates the absolute value MV* vector control commands output voltage of the inverter as a ratio to the maximum voltage VMmax (given by formula (9)), which can be output by the inverter. For example, in the case of PMF=1,0, the absolute value of VM* vector control commands output voltage of the inverter is equal to the maximum voltage VMmax, which can be output by the inverter.

As can be understood from formulas (2)to(10), the coefficient modulation PMF has the characteristic that the ratio PMF modulation changes according to the command id* current control d-axis and to the Mande iq* of the current control q-axis formed through the block 10 forming teams the current control.

Unit 40 calculation of governors of the phase angles calculates on the basis of the following formula (11) control phase angle θ of the commands vd* control voltage d-axis, which is the sum of the error pde current d-axis and forward voltage vdFF d-axis command vq* control q-axis voltage, which is the sum of the error pqe current q-axis and forward voltage vqFF q-axis, and the reference phase angle θe.

θ=θe+π+THV(11)

THV in the formula (11) is represented by the following formula:

THV=tan-1(vd*/vq*)(12).

The configuration and operation of the unit 50 a signal pulse-width modulation/control voltage is explained in relation to figure 3. Figure 3 is a diagram of an example of a detailed block configuration 50 a signal pulse-width modulation/control voltage, shown in figure 1.

As shown in figure 3, the block 50 a signal pulse-width modulation/control voltage includes a block 71 calculation of the signal pulse suppression, which receives the voltage is their EFC on the capacitor as input and generates a signal BTPMFCMP pulse suppression. The block 50 a signal pulse-width modulation/control voltage multiplying factor PMF modulation signal BTPMFCMP suppression of ripples to form PMFM, which is a command signal to control the amplitude of a control voltage. Configuration block 71 calculation of the signal pulse suppression is illustrated below.

Block 55 calculation control voltage forms on the basis of the following formula command Vu* control voltage of U-phase command Vv* control voltage V-phase and the command Vw* control voltage of the W-phase, which are commands to control three-phase voltage from the signal PMFM and the control phase angle θ.

Vu*=PMFM•sin θ(13)
Vv*=PMFM•sin(θ-(2•π/3))(14)
Vw*=PMFM•sin(θ-(4•π/3))(15)

The absolute value of the command Vu* control voltage of U-phase command Vv* control voltage V-phase and command Vw* control voltage of the W-phase, formed by block 55 calculation control voltage is compared with the carrier signal CAR through modules 61-63 comparison. Are formed the gate signals U, V and W and invert rowanne Gating signals X, Y and Z, inverted through the inverting circuits 64-66.

Carrier signal CAR is one of the signals selected in the switch 59 selecting by processor 60 pulse mode, acting as the switch unit pulse mode. Any of the asynchronous multipulse (in General, approximately 1 kHz) carrier signal A, generated by block 57 the formation of an asynchronous multi-pulse carrier signal, synchronous trichinoscope carrier signal B generated by block 58 forming synchronous credimpulso carrier, and zero values of C chosen in simultaneous single-pulse mode is selected through the switch 59 of choice. Asynchronous multi-pulse carrier signal and A synchronous trichinosis carrier signal B take values from -1 to 1, centered around zero.

The processor 60 of the switching pulse mode switches the switch 59 of choice according to the values of the coefficient modulation PMF and the control phase angle θ. In particular, in the region in which the ratio PMF modulation is low (equal to or less than 0,785), the switch 59 selection switches on the side of the asynchronous multi-pulse carrier signal A to select asynchronous multipulse mode. When the ratio PMF modulation more 0,785 and less than 1.0, the switch 59 in the boron switches on the side synchronous trichinoscope carrier signal B to select the synchronous pulse mode. When the ratio PMF modulation reaches approximately 1.0 (ratio PMF modulation can be 0,99 etc. and not only 1.0), the switch 59 selection switches on the side of the zero value C. In this configuration, at the time when the ratio PMF modulation equal to approximately 1.0, you can automatically switch to pulse mode to the synchronous single-pulse mode. On the contrary, when the ratio PMF modulation is less than approximately 1.0, you can automatically switch to pulse mode to the synchronous trichinosis mode. In other words, you can easily make the transition of the output voltage of the inverter 2 from minimum to maximum.

When switching pulse mode signal accessed by the processor 60 of the switching pulse mode, preferably is a factor PMF modulation, which is the signal before the signal BTPMFCMP pulse suppression, illustrated below, is reflected. By performing the configuration in which the appeal to the factor PMF modulation, it is possible to prevent unstable operation of the switching pulse mode by the processor 60 of the switching pulse mode.

Synchronous trichinosis mode pulse mode is required for the output voltage has a factor PMF modulation, equal to or greater than 0,785 that is not moativation in asynchronous multi-pulse mode. If method overmodulation is used in asynchronous multi-pulse mode, synchronous pathinalam mode, synchronous devotionalism mode and so on, you can output a voltage equivalent to the voltage in the synchronous Trichinella mode. However, when this method is used, the ratio PMF modulation and the output voltage of the inverter 2 are extremely nonlinear. Therefore, it is necessary to correct for this nonlinearity. There is a disadvantage in that the configuration is complicated.

In the above explanation, the threshold factor PMF modulation for switching asynchronous multi-pulse carrier signal and synchronous trichinoscope carrier signal is set equal 0,785. However, you may use a threshold value that is different from 0,785.

As explained below, the carrier signal CAR compared with the control voltage is at least asynchronous multi-pulse carrier signal and a synchronous carrier signal. Carrier signal CAR can be chosen according to the pulse mode is selected by processor 60 pulse mode, acting as the control unit of the pulse mode.

Asynchronous multi-pulse carrier signal is the carrier signal, determining the frequency regardless of the output frequency FINV of the inverter. Frequency sostavlyayushchaya 1000 Hz.

Frequency synchronous carrier signal, such as a synchronous trichinosis carrier signal is determined as a function of output frequency FINV of the inverter so that the number of pulses and position pulses included in the output voltage of the inverter are identical in half the positive side and the negative half-cycle of the output side voltage of the inverter. In this embodiment explains an example in which only synchronous trichinosis carrier signal is used as a synchronous carrier signal. However, synchronous carrier signal may be, for example, synchronous pathinalam bearing signal and the like, in addition to synchronous trichinoscope carrier signal. Many synchronous load-bearing signals may be prepared and included as needed.

In a state where it is selected asynchronous multipulse mode when the output frequency FINV of the inverter is located near the frequency of the asynchronous multi-pulse carrier signal with a reduced number of pulses included in a half-period of the output voltage of the inverter. Frequency asynchronous multi-pulse carrier signal is a value defined regardless of the output frequency FINV of the inverter. Therefore, when the motor 6 is excited in this state, the number impulse and position pulses, accordingly included in the positive half-cycle and a negative half-cycle of the output voltage of the inverter, are derived from the equilibrium or temporarily fluctuate, and positive and negative symmetry of the voltage applied to the motor 6, is broken. The oscillation current and torque pulsations occur in the electric motor 6, which causes noise and vibration.

On the other hand, when using synchronous carrier signal, the number of pulses and position pulses, respectively, included in the positive half-cycle and a negative half-cycle of the output voltage of the inverter is equal, and is provided with positive and negative symmetry of the voltage applied to the motor 6. Consequently, it is possible to prevent the occurrence of fluctuations in the current and torque pulsations in the motor 6 and steadily to excite the motor 6.

Regarding simultaneous single-pulse mode the number of pulses included in a half-period of the output voltage of the inverter, always equal to one and is fixed without the temporary changes. Therefore, the number of pulses and position pulses are identical in the positive half-cycle and a negative half-cycle of the output voltage of the inverter. Can be provided with positive and negative symmetry h is voltage, applied to the motor 6. Therefore, there is no concern that the oscillation current and torque pulsations occur in the motor 6.

Can be added configuration for precise control of the timing switch to pulse mode according to the Manager of the phase angle θ. Provided such an advantage that it is possible to suppress the pulsation of the motor current during the switching pulse mode.

The configuration and operation of the block 71 calculation of the signal pulse suppression, which is acting as a signal pulse suppression, explained in relation to figure 4. Figure 4 is a diagram of an example of a detailed block configuration of 71 calculation of signal suppression pulse, shown in figure 3.

In block 71 calculation of the signal pulse suppression, as shown in figure 4, the voltage EFC on the condenser is introduced into the band-pass filter (hereinafter referred to as "BPF") 72. The voltage EFC on the capacitor is filtered by BPF 72, and a signal is generated EFCBP1. BPF 72 is set such that 2f-component frequency power supply source 230 AC power can be efficiently retrieved.

In the adder 73, a signal is generated EFCBP2 as the sum of the generated signal EFCBP1 and command EFC* control voltage on the capacitor, which is a command to control the population by the voltage of the capacitor 1. Command EFC* control voltage on the capacitor is the target value of the voltage EFC on the capacitor when the Converter 220 performs control to convert an AC voltage source 230 AC power into a DC voltage (=voltage EFC on the capacitor). Usually the command EFC* control of the capacitor voltage takes a value of approximately 1500-3000 Century

Instead of the command EFC* control voltage on the capacitor may be formed from a signal obtained by passing the voltage EFC on the capacitor through the LPF (not shown) and remove the variable component of the current so as to leave only the DC component of the current. This signal may be added to the signal EFCBP1 by the adder 73 to form the signal EFCBP2.

Command EFC* control voltage on the capacitor and the signal EFCBP2, which is an output of the adder 73, entered into the divider 74. In the divider 74 command EFC* control voltage on the capacitor divided by the signal EFCBP2. The result of the division is output as the signal BTPMFCMP suppression of ripples.

Similarly, the signal EFCBP2 instead of the command EFC* control voltage on the capacitor may be formed from a signal obtained by passing the voltage EFC on the capacitor through the LPF (not shown) and delete the variable stood at the gates of the current to leave only the DC component of the current. This signal can be divided into signal EFCBP2 through divider 74 to form the signal BTPMFCMP suppression of ripples.

The signal BTPMFCMP pulse suppression, thus obtained, indicates the inverse number of the ratio of the voltage EFCBP2 on the capacitor, which includes a ripple component 2f-component of the power source to the direct current component of the voltage EFC on the capacitor.

Figure 5 is a chart example of the internal state of the block 71 calculation of the signal pulse suppression in the first embodiment. Figure 5 internal state, in which the Central value of the voltage EFC on the capacitor is 3000, shown as an example.

As shown in figure 5, the voltage EFC on the capacitor includes, together with 2f-component of the power supply ripple component generated by the operation of the switching Converter 220 and having a frequency above the frequency of 2f-component of the power source (see the wave shape at the top of the stepped part of the figure).

The signal EFCBP1 is a signal from which the ripple component is removed by the BPF 72, and includes only the 2f component of the power source (see the wave shape in the middle upper part of the drawing).

The signal EFCBP is the value obtained by adding EFC*, which is a control command to the voltage on the capacitor to the signal EFCBP1. Only 2f-component of the power source is included in the signal EFCBP2 as a component fluctuations (see wave shape in the middle of the lower stepped part of the figure).

You can see that the signal BTPMFCMP suppress pulsations indicates the inverse voltage EFCBP2 on the capacitor, which includes a ripple component 2f-component of the power source, relative to the direct current component of the voltage EFC on the capacitor (see wave shape in the middle of the lower stepped part of the figure).

The signal BTPMFCMP pulse suppression, which is an output of the block 71 calculation of the signal pulse suppression, is injected into the multiplier 70 unit 50 a signal pulse-width modulation/control voltage and multiplied by a factor PMF modulation (see figure 3). By multiplying the signal BTPMFCMP pulse suppression by a factor PMF modulation can be used to generate signal PMFM commands control the amplitude of a control voltage to suppress the ripple component 2f-component of the power source voltage EFC on the capacitor.

As shown in figure 3, the control commands output voltage of the inverter 2 are formed on the basis of the signal PMFM team upravleniemoeda control voltage. Thus, it is possible to adjust the width of the voltage pulse output by the inverter to suppress 2f-component of the power source. Consequently, it is possible to solve the problem that the AC motor generates a current overload, and too much torque pulsations occur in the region in which the output frequency FINV of the inverter and the frequency of 2f-component of the power source are located close to each other.

The configuration and the operation unit 80, the control commands of the current control is illustrated in relation to 6. 6 is a diagram of an example of a detailed block configuration 80 of the control commands control the current, shown in figure 1.

The block 80 of the control commands of the current control is a component having the function of forming on the basis of the output frequency FINV of the inverter, the regulating values dV command current control. The block 80 of the control commands of the control current includes, as shown in Fig.6, the block 85 generate control commands modulation depth, the block 84 subtraction unit 81 limitations and amplifier 82 (having a gain K).

Block 85 generate control commands modulation factor works as a unit forming task teams amplitude control voltage and generates on the basis of the output frequency FINV of the inverter command PMF* control the value of the modulation depth, which is the target command amplitude control voltage. Unit 84 outputs the subtraction value obtained by subtracting the ratio PMF modulation of command PMF* control coefficient modulation. Block 81 constraint takes the output of block 84 subtraction as an input signal. When the sign of the input signal is positive, the block 81 limit sets the output equal to zero. When the sign of the input signal is negative, the block 81 restrictions directly outputs the input signal. The amplifier 82 (having a gain K) amplifies the output signal and outputs the amplified signal as a regulating value dV command current control. Regulating the value of the dV command current control is presented, as indicated by the following formula:

dV=LIM(PMF*PMF)•K(16),

where LIM() represents a function for limiting the upper and lower limit values in parentheses according to the method explained above.

The configuration and operation of the block 85 generate control commands modulation depth is illustrated in Fig.7 relation. Fig.7 is an enlarged diagram of the block 85 generate control commands modulation depth, is shown in Fig.6.

As explained above, the block 85 the formation of management teams is oefficients modulation generates from the output frequency FINV of the input inverter command PMF* control coefficient modulation. As shown in Fig.7, the command PMF* control the modulation depth is set to, for example, equal to 0.95 in the region in which the output frequency FINV of the inverter is set within 120 Hz (120 Hz ± 30 Hz), and is set equal to 1.0 in other areas.

Through the configuration of the block 85 generate control commands modulation factor of thus, when the output frequency FINV of the inverter is approximately 120 Hz (120 Hz ± 30 Hz), it is possible to create and manage regulatory value dV command current control so that the ratio PMF modulation is 0.95.

The area in which the ratio PMF modulation is set to 0.95, and is illustrated as the area in which the output frequency FINV of the inverter is set within 120 Hz (120 Hz ± 30 Hz). However, this is an example in which the frequency source 230 AC power is 60 Hz. This is due to the fact that 120 Hz is equivalent to 2f-component of 60 Hz. On the other hand, when the frequency source 230 AC 50 Hz, because the 2f component is 100 Hz, the region in which the ratio PMF modulation is set to 0.95, and is the area in which the output frequency FINV of the inverter is 100 Hz (100 Hz ± 30 Hz).

When the configuration on the basis of 6 and 7, at the point when the ratio PMF modulation exceeds a pre-defined is nnow command PMF* control the modulation depth, the input signal in block 81 limit is reduced to be equal to or less than zero, and may be formed of a negative regulatory value dV command current control. Consequently, it is possible to control the magnetic flux weakening to ensure coincidence of the output voltage of the inverter 2 to the value specified by the command PMF* control the modulation depth.

In particular, when the command voltage is acceptable margin relative to the maximum output voltage of the inverter 2, which regulates the value of the dV command current control is not displayed. At the point when the ratio PMF modulation exceeds the command PMF* control the modulation depth (the point when the command control voltage exceeds the maximum voltage specified by the command PMF* control coefficient modulation), a negative value occurs in the output unit 81 restrictions, and output regulating value dV command current control. Therefore, an optional current id d-axis current is not supplied and the electric current of the electric motor 6 can be minimized.

Fig is a diagram to explain the relationship between output frequency FINV of the inverter and the transition factor PMF modulation, the transition pulse mode and the transition of the switch 59 of choice (see IG) in the first embodiment. The acceleration of the vehicle with an electric motor from a stopped condition is illustrated in the example below.

As shown in Fig when the vehicle motor is moving at low speed, i.e. when the output frequency FINV of the inverter is low, the ratio PMF modulation is small, the pulse mode is an asynchronous multi-pulse mode, and A is selected as the switch 59 of choice. On the other hand, when the speed of the vehicle, the electric motor is increased, and the ratio PMF modulation is increased to be equal to or greater 0,785, the output voltage of the inverter 2 is saturated in asynchronous multi-pulse mode. Therefore, the switch 59 selection switches to B, and pulse mode switches to synchronous trichinosis mode. When the speed of the vehicle, the electric motor is additionally increased, and the ratio PMF modulation reaches 1.0, the switch 59 of the selection switches in C, and the pulse mode is switched to the synchronous single-pulse mode.

The deceleration of the vehicle with the electric motor through the use of regenerative braking is not shown in the drawing. However, according to the order opposite to the order explained above, the pulse mode is transferred from the synchro is nogo single-pulse mode to the synchronous trichinosis mode and asynchronous multi-pulse mode. The switch 59 of the selection switches in the order C, B and A.

Advantages of the NAT device and power for excitation of the motor according to this variant of the implementation are explained relative to control operations for components, explained above.

Fig.9 is a graph of total characteristics control of synchronous motors with permanent magnets in the first embodiment of the present invention and the example in the prior art. Control characteristics shown in Fig.9 are the characteristics of the control relative to synchronous motors with permanent magnets, designed for vehicles with an electric motor. It is assumed that the maximum output torque of 1,500 Newton metres, and the input voltage EFC is 3000 C. Synchronous motor with permanent magnets that operate according to other parameters, allows similar characteristics.

Figure 9, the abscissa represents the current id d-axis, and the ordinate represents the current iq of the q-axis. A set of curves (solid lines)present from the area in the top right to the bottom left in the drawing, are fixed torque curves are curves indicating the relationship between current id d-axis current iq of the q-axis when the corresponding torque of the PTO shall nth T, described in the left-hand end of the drawing (the ratio between the current vectors). Curve (dotted line) from the area in the top left to the bottom right in the drawing is a curve indicating the minimum current, and is the curve which minimizes the motor current, displayed when a certain torque T. in Other words, the curve is a curve that indicates the mode in which it is possible so-called control maximum torque/current for the formation of maximum torque with minimum current.

If the vector current is controlled to the point of intersection of the curve indicating the minimum current, and a fixed torque curve, you can get the torque T with the minimum current. By performing such a control is provided such an advantage that it is possible to minimize losses in the winding and losses in the inverter of the electric motor 6 when you get a certain torque T, and you can reduce the size and weight of the electric motor 6 and the inverter 2. For example, when the required output torque T of 1000 Newton metres, if the control current is performed by the inverter 2 so that the current id d-axis is close to -125 and the current iq of the q-axis is close to 225 a (point P1, as shown in the drawing), 1000 Newton-meters can be formed by means of the CMV minimal current.

Drawing curves (lines with alternate long and short dash), drawn in the form of hills, are the limiting voltage curves, which are fixed curves induced voltage, and are curves indicating the relationship between current id d-axis current iq of the q-axis, at which the voltage across contact terminals of the electric motor 6 is maximized at a certain output frequency FINV of the inverter (the ratio between the current vectors). The drawing shows the limit curves of the voltage in five cases (60 Hz, 90 Hz, 120 Hz, 150 Hz and 180 Hz), in which the output frequency FINV of the inverter is set as the parameter, provided that the input voltage EFC of the inverter 2 is set to 3000 C.

The combination of current id d-axis current iq of the q-axis, which can logically be selected (vector-current), is located on the inner side of the marginal curves voltage (the bottom curve). When the motor 6 works with vectors of the current present on the line limit curves of voltage, line voltage of the motor 6 is maximized (i.e. the state in which the ratio PMF modulation of the inverter 2 is 1.0, and the maximum output voltage). Torque T that can be displayed on the stage is a torque T at the point of intersection of the limiting voltage curve and a fixed torque curve is amenta.

When the motor 6 works with vectors of the current present on the inner side (lower side) of the limiting curves of voltage, line voltage of the motor 6 takes a value equal to or greater than zero and less than the maximum value (i.e. the ratio PMF modulation inverter 2 less than 1.0).

The vector current is present on the outer side of the marginal curves voltage (top curves), can't get out, because the current vectors are in an area that exceeds the maximum output voltage of the inverter 2.

As you can see from the limiting curves of voltage when the output frequency of the inverter FINV (60 Hz, 90 Hz, 120 Hz, 150 Hz and 180 Hz) in these five cases shown in figure 9, as the speed of the motor 6 increases and the output frequency FINV of the inverter increases, the voltage limit curves are moved to the lower side of the drawing, the current vectors which can be selected are limited, and the maximum value of torque T that can be displayed is reduced. As the output frequency FINV of the inverter increases, the torque T that can be formed on a curve indicating the minimum current also decreases.

When the voltage EFC on the capacitor increases, the marginal curve of the voltage at the identical output frequency FIV inverter is moved to the upper side in the drawing. When the voltage EFC on the capacitor decreases, the marginal curve of the voltage at the identical output frequency FINV of the inverter is moved to the lower side in the drawing.

For example, when the output frequency FINV of the inverter is 60 Hz, operating point that satisfies the minimum current at maximum torque of 1,500 Newton metres (about current id d-axis = -175 A, about current iq of the q-axis = 295 A, point A in the drawing), is a point sufficiently remote to the lower side from the limit voltage curve.

On the other hand, when the output frequency FINV of the inverter is 150 Hz, the maximum torque that can be generated is approximately 1200 Newton-meters (point P2 in the drawing) about current id d-axis = -245 A and about current iq of the q-axis = 200 A at maximum voltage curve. Similarly, the maximum torque that can be generated in mode minimum current of approximately 930 Newton-meters (point P3 in the drawing) about current id d-axis = -120 A and about current iq of the q-axis = 210 A to limit voltage curve. Mode minimum current is impossible in the field from 930 nm up to 1200 meters Newton-meters. This is the area in which work is possible through the implementation of the control magnetic flux weakening for negative increasing the current id d-axis.

However, as the management mA the magnetic flux weakening is deeper (as the current id d-axis negative increases), the vector current generated by the current id d-axis current iq of the q-axis increases, and the electric current of the electric motor 6 increases.

In particular, in order to minimize losses in the motor winding 6 and losses of the inverter 2, it is desirable to control the inverter 2 so as to select the vector current (a combination of current id d-axis current iq of the q-axis) on the curve mode the minimum voltage to the maximum extent possible and to instruct the motor 6 to generate the desired torque. When the output frequency FINV of the inverter is increased according to the increase in the speed of rotation of the motor 6, in the area in which the desired torque cannot be output on the curve mode minimum current due to restrictions limiting voltage curve, in General, the current id d-axis increases negatively, and to control the magnetic flux weakening.

In addition to the control mode minimum current, explained above (control the maximum torque/current), you can also control vector current on the curve of maximum efficiency (not shown), for which the loss of the electric motor 6, which includes the core losses of the motor 6, minimized, and apply the management on the basis of maximum efficiency for motor control 6.

Explains the working features and advantages of the ka to the present invention, performed in the area in which the control switches to synchronous optical mode (i.e. the region in which the ratio PMF modulation takes a value close to 1.0) or when the output frequency FINV of the inverter is close 2f-component of the power source during operation in a synchronous optical mode.

First, explain the operation of the control in the above example, in order to clarify the details of the problem. Then the means of resolving problems in the first embodiment of the present invention is illustrated in relation to Fig. Fig is a chart to explain the operation mode in the example in the prior art, on which MPM, 3PM and 1PM mean asynchronous multipulse mode, synchronous trichinosis mode and simultaneous single-pulse mode, respectively. An example of control in which the motor 6 is started and accelerated from a state in which the motor 6 is stopped is displayed on Fig. Operating points A, B, C1, D, and E, shown in Fig respectively, correspond to the operating points A, B, C1, D, and E shown in Fig.9.

On Fig, first, the inverter 2 is started at the zero of the time scale, the command torque T is set equal to 1500 Newton meters, and voltage is applied to the motor 6 to start acceleration. Here the factor PMF modulation Uwe is aceveda from zero in proportion to the output frequency FINV of the inverter. Until the ratio PMF modulation does not reach 0,785 from zero of the time scale, asynchronous multi-pulse mode is selected as the switching mode of the inverter 2, and the torque T is fixed at 1500 Newton-metres. Therefore, the motor 6 is linearly accelerated, and the output frequency FINV of the inverter increases linearly.

At the point when the modulation depth reaches 0,785, the pulse mode is switched to the synchronous trichinosis mode. Between point a and point B, as the ratio PMF modulation reaches its maximum value of 1.0, the pulse mode is switched from the synchronous trichinoscope mode in synchronous single-pulse mode. Between point A (the output frequency of the inverter FINV = 60 Hz) and point B (the output frequency of the inverter FINV = 90 Hz), the command torque T decreases from 1500 Newton meters in inverse proportion to the output frequency FINV of the inverter. After the rate modulation PMF reaches 1,0, formed the magnitude of the dV regulation control current increases to the negative side according to the increase of the output frequency FINV of the inverter. Therefore, because the command id* current control d-axis negative increases, to control the magnetic flux weakening. Therefore, the command id* current control d-axis adjusted so that the ratio PMF modulation coincides with the command PMF* control modulation depth (=1,0).

Path vector current control mode explained above is explained in relation to figure 9. Figure 9, as explained above, since the working point A is on the lower side limit voltage curve and away from the limit voltage curve, the ratio PMF modulation of less than 1.0, and the output voltage of the inverter 2 is one that is less than the maximum voltage that can be output.

At operating point B, the command torque T is 1400 Newton-meters, and the vector current is controlled to the point at which the command id* current control d-axis is approximately -170 and command iq* of the current control q-axis is approximately 277 A. In this operating point B, the vector current is also supported on the limit curve of the voltage at FINV = 90 Hz. Regulating the value of the dV command current control is formed and operated in such a way that the ratio PMF modulation equal to 1.0.

At this operating point C1, the command torque T is 1200 Newton-meters, and the vector current is controlled to the point at which the command id* current control d-axis is -160 and command iq* of the current control q-axis is approximately 243 A. In this operating point C1 vector current is also supported on the limit curve of the voltage at FINV = 120 Hz. Regulating the value of the dV command current control is formed and operated in such a way that efficient PMF modulation equal to 1.0.

In the operating point D, the command torque T is 1100 Newton metres, and the vector current is controlled to the point at which the command id* current control d-axis is -177, and command iq* of the current control q-axis is approximately 220 A. In this operating point D, the vector current is also supported on the limit curve of the voltage at FINV = 150 Hz. Regulating the value of the dV command current control is formed and operated in such a way that the ratio PMF modulation equal to 1.0.

In the operating point E, the command torque T is 1000 Newton-meters, and the vector current is controlled to the point at which the command id* current control d-axis is -180 and command iq* of the current control q-axis is approximately 195 A. In this operating point E, the vector current is also supported on the limit curve of the voltage at FINV = 180 Hz. Regulating the value of the dV command current control is formed and operated in such a way that the ratio PMF modulation equal to 1.0.

Thus, in the example of control in prior art management, the working point moves from the operating point A to operating point B, C1, D and E. After the rate modulation PMF reaches 1,0, regulating the value of the dV command current control is formed to maintain the output voltage of the inverter 2 at the maximum value that can be displayed is ri simultaneous output torque T (in order to maintain a ratio PMF modulation = 1,0). The control magnetic flux weakening is performed according to the command id* current control d-axis, which includes the value of the dV regulation team of the current control.

According to management, the example of control in the prior art after the modulation depth reaches 1,0 to maintain a ratio PMF modulation equal to 1.0 and to maintain the applied voltage to the AC motor at the maximum, simultaneous single-pulse mode is selected in the switching circuit of the inverter. In the workspace, this simultaneous single-pulse mode, as explained above, since the regulation of the pulse width cannot be executed, the control for suppressing 2f-component of the power source can not be done, in particular, in the region in which the output frequency FINV of the inverter is located near 2f-component of the power source. Therefore, a problem arises in that the AC motor generates a current overload, and there is too much torque pulsations.

The operation control in the first embodiment, to solve the problem explained in relation to figure 10. Figure 10 is a diagram for explanation of the control modes in the first embodiment of the present invention. Shows an example of control, done is imago, when the motor 6 is started and accelerated from a state in which the motor 6 is stopped. Operating points A, B, C, D and E, shown in figure 10, respectively, correspond to the operating points A, B, C, D and E shown in Fig.9.

Figure 10, first, the inverter 2 is started at the zero of the time scale, the command torque T is set equal to 1500 Newton meters, and voltage is applied to the motor 6 to start acceleration. Factor PMF modulation increases from zero in proportion to the output frequency FINV of the inverter. Until the ratio PMF modulation does not reach 0,785 from zero of the time scale, asynchronous multi-pulse mode is selected as the switching mode of the inverter 2, and the torque T is fixed at 1500 Newton-metres. Therefore, the motor 6 is linearly accelerated, and the output frequency FINV of the inverter increases linearly.

At the point when the modulation depth reaches 0,785, the pulse mode is switched to the synchronous trichinosis mode. Between point a and point B, as the ratio PMF modulation reaches its maximum value of 1.0, the pulse mode is switched from the synchronous trichinoscope mode in synchronous single-pulse mode. Between point A (the output frequency of the inverter FINV = 60 Hz) and point B (the output frequency of the inverter FINV = 90 Hz), the command torque Tumenskaya from 1500 Newton meters in inverse proportion to the output frequency FINV of the inverter. After the rate modulation PMF reaches 1,0, formed the magnitude of the dV regulation control commands negative current increases according to the increase of the output frequency FINV of the inverter. Therefore, because the command id* current control d-axis negative increases, to control the magnetic flux weakening. Therefore, the command id* current control d-axis adjusted so that the ratio PMF modulation coincides with the command PMF* control modulation depth (=1,0). The above control operation is equivalent to the operation in the example in the prior art.

On the other hand, the region between the working point B and the working point D is the region in which the output frequency FINV of the inverter and 2f-component of the power source are located near each other. The working point C is a point at which the output frequency FINV of the inverter is 120 Hz, and is the point at which the frequency of 2f-component of the power source when the frequency source 230 AC power is 60 Hz, and the output frequency FINV of the inverter accurately coincide with each other.

Therefore, in this embodiment, in the range of the operating point B to the operating point D, which is the range in which the output frequency FINV of the inverter and 2f with the other commercial power source are located close to each other or coincide with each other, command PMF* control the modulation depth decreases from 1.0 to 0.95. According to this control deviation between factor PMF modulation and command PMF* control coefficient modulation. Therefore, regulating the value of the dV command current control is formed on the basis of this deviation. Formed command id* current control d-axis works on the basis of the regulating values dV command current control, to further increase in the negative direction. Therefore, the command id* current control d-axis and command iq* of the current control q-axis are formed as a vector of the current located at a fixed torque curve corresponding to the command torque T, and on the inner side (lower side) of the limiting voltage curve around FINV = 120 Hz. According to the command id* current control d-axis and the command iq* of the current control q-axis formed thereby control the magnetic flux weakening applied to the motor 6 becomes deeper, and the induced voltage of the motor 6 additional drops. Therefore, the ratio PMF modulation decreases as well. Factor PMF modulation is controlled so as to coincide with the command PMF* control the modulation depth.

In this region the device power conversion for excitation e is extradigital according to this variant implementation reduces the coefficient modulation PMF so, so it was less than usual factor modulation to switch to pulse mode in synchronous trichinosis mode, which is a synchronous pulse mode. Therefore, according to the signal BTPMFCMP pulse suppression, which is an output of the block 71 calculation of the signal pulse suppression, it is possible to adjust the pulse width for the output voltage outputted by the inverter 2. You can manage to suppress 2f-component of the power source. According to this control, you can solve the problem that the AC motor generates a current overload occurs excessively large torque pulsations.

Because synchronous trichinosis mode, which is a synchronous pulse mode, is selected as the pulse mode, the number of pulses and position pulses, respectively, included in the positive half-cycle and a negative half-cycle of the inverter 2, equal, and is provided with positive and negative symmetry of the voltage applied to the motor 6. Consequently, it is possible to prevent the occurrence of fluctuations in the current and torque pulsations in the motor 6, to prevent the occurrence of noise and fluctuations due to fluctuations in current and torque pulsations and perform stabil the Noah excitation of the motor 6. Operations after operating point D are identical to the operations in the example in the prior art explained above.

Thus, when the control according to this variant implementation, the working point moves in the order of the operating points A, B, C, D and E. the Command torque T at operating points A, B, C, D and E, respectively, are 1500 Newton meters, 1400 Newton-meters, 1200 Newton-meters, 1100 Newton metres and 1000 Newton meters. The command torque T are identical to the commands in the operating points A, B, C1, D and E of example in the prior art. In other words, in this embodiment, at the time when the control for suppressing 2f-component of the power supply occurs, the output torque T is not affected in operating points, which includes operating point C.

On the other hand, since the working point C1 in the example in the prior art and the working point C in this embodiment are identical to the fixed torque curve (at 1200 Newton-metres), corresponding to a predefined value of the command torque control, the torque output by an electric motor 6, is the same as in the working point C1, and the operating point C. in Other words, in this embodiment, the can is about to reduce the coefficient modulation PMF, for example, to 0.95, while maintaining the output torque of the electric motor 6 unchanged, reducing the induced voltage of the motor 6 and lowering the output voltage of the inverter 2. Because the ratio PMF modulation is reduced to be lower than the coefficient of modulation, and pulse mode switches to synchronous trichinosis mode, which is a synchronous pulse mode, you can adjust the width of the voltage pulse output by the inverter 2 according to the output unit 71 calculation of the signal pulse suppression and manage to suppress 2f-component of the power source. Therefore, resolves the problem in the prior art in that the AC motor generates a current overload occurs excessively large torque pulsations.

Generated control command current command id* current control d-axis and command iq* of the current control q-axis), in which the ratio PMF modulation coincides with the command PMF* control coefficient modulation. Consequently, it is possible to reduce the ratio PMF modulation, for example, to 0.95 while maintaining the output torque of the electric motor 6 when the pre-defined value, reducing the induced voltage of the motor 6 and the pony is th output voltage of the inverter 2.

In the example explained above, the motor 6 is accelerated from a stopped state. However, the configuration explained in this embodiment can also be applied when the motor 6 is subjected to the operation of recovery and stops during high-speed rotation.

The second option exercise

In the first embodiment, is disclosed a configuration in which the control value of the control current to the control command of the current control device for converting power for excitation of the motor is controlled properly, or switching pulse mode is managed properly to provide for the regulation of the pulse width of the voltage outputted by the inverter 2, and to ensure the efficient execution control to suppress 2f-component of the power source included in the output voltage of the inverter 2. In the second embodiment, is disclosed a configuration in which a control command voltage of the inverter to control the inverter 220 is additionally formed properly to ensure effective reduction of the electric current supplied to the motor 6.

11 is a diagram of a configuration example of the NAT device power dawsbergen motor according to the second variant of implementation of the present invention. A more detailed configuration of the inverter 220, which is the first conversion unit capacity, in the configuration of the NAT device and power for excitation of the motor shown in figure 1, explained below. Of the components shown figure 11, the components identical to the components shown in 11, already explained. Therefore, mainly explains the components associated with the second option implementation.

As shown in figure 11, the ratio PMF modulation and the output frequency FINV of the inverter formed by the second unit 100 controls the voltage EFC on the capacitor is determined by a sensor 8 voltage, and input current IS determined by a sensor 214 current introduced into the first control unit 200. This first control unit 200 is a component having a function to control the output voltage (DC voltage) Converter 220, and includes a block 210 generate control commands DC voltage and unit 280 controls the DC voltage.

Block 210 generate control commands DC voltage generates a command EFC* control DC voltage, which is the target value of the voltage on the capacitor and is the command EFC* control voltage on the capacitor. Nl is to 211 control accepts input voltage command EFC* control the DC voltage and the voltage EFC on the capacitor and generates on the basis of the deviation between the command EFC* control DC voltage and the voltage EFC condenser team IS* manage current and outputs a command IS* current control. Block 212 current control accepts input commands IS* manage current and the input current IS determined by a sensor 214 current, and generates on the basis of the deviation between the command IS* manage current and input current IS VC* control voltage transformer. Block 213 of the signal pulse-width modulation takes command VC* control voltage of the Converter and generates a signal on/off (pulse width modulation) CG for the switching element (not shown) of the inverter 220 to ensure coincidence of the voltage on the input side (the side of the power source AC) Converter 220 with VC* control voltage of the Converter.

Using functions unit 211 of the control voltage, block 212, the current control unit 213 of the signal pulse-width modulation, configured as explained above, the unit 280 controls the DC voltage generates a signal pulse-width modulation CG using command EFC* control voltage DC voltage EFC on the capacitor and the input current IS, and outputs a pulse width modulation CG in the Converter 220.

Detailed configuration of the operation unit 210 to generate control commands DC voltage is illustrated below in relation to Fig. Fig is a diagram of a first example of the configuration of the block 210 to generate control commands DC voltage in the second embodiment, shown in 11.

As shown in Fig, block 210 generate control commands DC voltage, which is the first configuration example includes a table 240 commands control the DC voltage. Table 240 commands control the DC voltage generates, based on the output frequency FINV of the inverter command EFC* control DC voltage and outputs a command EFC* control DC voltage.

When the output frequency FINV of the inverter is not about frequency 2f-component of the power source, the table 240 commands control the DC voltage outputs voltage within the estimated time as command EFC* control DC voltage. On the other hand, when the output frequency FINV of the inverter is located near the frequency 2f-component of the power source, the table 240 commands control the DC voltage outputs a command of the control voltage over voltage over the estimated time command EFC* control DC voltage.

For example, when the frequency of the power source AC is 60 Hz, if the output frequency is the frequency FINV of the inverter is not in the range of about 90-150 Hz, centered around 120 Hz, which is the frequency of 2f-component of the power source, the table 240 commands control the DC voltage outputs, for example, 3000 In as command EFC* control DC voltage. If the output frequency FINV of the inverter is in the range of about 90-150 Hz, table 240 commands control the DC voltage outputs, for example, 3300 V, which is obtained by increasing the voltage over the estimated time 5-10%, as the command EFC* control DC voltage.

By the configuration unit 210 generate control commands DC voltage thus, when the output frequency FINV of the inverter is in the region close to the frequency 2f-component of the power source, for example, the field of 90-150 Hz, it is possible to control the voltage EFC on the capacitor so that it is high, and, therefore, increase the maximum voltage that can be output by the inverter 2. According to this control, you can reduce the required value of the magnetic flux weakening. As a result, you can decrease the value of the dV regulation control current and to reduce the absolute value of the command id* current control d-axis. Consequently, it is possible to reduce the electric current of the electric motor 6 and cf is the ranking, with an electric current displayed when the configuration of the first control unit 200 according to the second variant of implementation not applicable.

Block 210 generate control commands DC voltage is not limited to the configuration shown in Fig, and may be configured, for example, as shown in Fig. Fig is a diagram of a second example of the configuration of the block 210 to generate control commands DC voltage in the second embodiment, shown in 11.

Block 210 generate control commands DC voltage shown in Fig, is a component that forms on the basis of the output frequency FINV of the inverter and command PMF* control rate modulation as a target value of the coefficient of PMF the modulation index amplitude of the voltage command EFC* control DC voltage and includes a table 250 control commands modulation depth, unit 251 subtraction, block 252 restrictions, block 253 proportional integration and the adder 254.

Table 250-control commands modulation factor of forms-based input output frequency FINV of the inverter, the command PMF* control coefficient modulation. Block 251 subtraction takes input factor PMF modulation and command PMF* control the modulation depth, generates a signal deviation, the floor is obtained by subtracting the command PMF* control of the modulation factor of the coefficient of modulation PMF, and outputs a signal deviations in block 252 restrictions. When the sign of the input signal is positive, the block 252 restrictions directly outputs the input signal. When the sign of the input signal is negative, the block 252 restrictions outputs zero regardless of the value of the input signal. Block 253 proportional integration outputs a value obtained by calculating the proportional integral signal output unit 252 restrictions. The adder 254 adds the output signal of block 253 proportional integration and team EFC0* control reference DC voltage (for example, 3000), and outputs the added signal as the command EFC* control DC voltage.

For example, when the frequency of the power source AC is 60 Hz, if the output frequency FINV of the inverter is not in the range of about 90-150 Hz, centered around 120 Hz, which is the frequency of 2f-component of the power source, the table 250 control commands modulation factor of displays, for example, of 1.0 as the command PMF* control coefficient modulation. On the other hand, if the output frequency FINV of the inverter is in the range of about 90-150 Hz, table 250 control commands modulation factor of displays, for example, of 0.95 as command PMF* control the modulation depth.

Group is a rotary configuration unit 210 generate control commands DC voltage thus when the output frequency FINV of the inverter is, for example, in the field of 90-150 Hz, you can increase the voltage EFC on the capacitor so that the ratio PMF modulation of the inverter 2 is, for example, of 0.95. Consequently, it is possible to increase the maximum voltage that can be outputted by the inverter 2. According to this control, you can reduce the required value of the magnetic flux weakening. As a result, you can decrease the value of the dV regulation control current and to reduce the absolute value of the command id* current control d-axis. Consequently, it is possible to reduce the electric current of the electric motor 6 as compared with an electric current that is output when not used, the configuration of the first control unit 200 according to the second variant implementation.

When the configuration of the first variant implementation explained above, it is possible to reduce the ratio PMF modulation, for example, to 0.95 while maintaining the output torque of the electric motor 6 when the pre-defined value, the control commands, reducing the induced voltage of the motor 6 and lowering the output voltage of the inverter 2 in the region in which the output frequency FINV of the inverter is located near the frequency 2f-component of the power source. Therefore, since the pulse mode is switched to SYN the electronic trichinosis mode, which is synchronous pulse mode, you can adjust the width of the voltage pulse output by the inverter 2 according to the signal BTPMFCMP pulse suppression, which is an output of the block 71 calculation of the signal pulse suppression, and manage to suppress 2f-component of the power source. Therefore, resolves the problem that the AC motor generates a current overload occurs excessively large torque pulsations.

When the configuration of the first variant implementation of the generated control command current command id* current control d-axis and command iq* of the current control q-axis), in which the ratio PMF modulation coincides with the command PMF* control coefficient modulation. Consequently, it is possible to reduce the ratio PMF modulation, for example, to 0.95 while maintaining the output torque of the electric motor 6 when the pre-defined value, reducing the induced voltage of the motor 6 and lowering the output voltage of the inverter 2.

When the configuration of the first variant implementation, because synchronous trichinosis mode, which is a synchronous pulse mode, is selected as the pulse mode, the number of pulses and position pulses, respectively, is included in will put the local half-cycle and a negative half-cycle of the inverter 2, equal, and is provided with positive and negative symmetry of the voltage applied to the motor 6. Consequently, it is possible to prevent the occurrence of fluctuations in the current and torque pulsations in the motor 6, to prevent the occurrence of noise and fluctuations due to fluctuations in current and torque pulsations and to perform stable excitation of the motor 6.

Additionally, when the configuration of the second variant implementation, the advantage of reducing the electric current of the electric motor 6 is large compared to the configuration of the first variant implementation, which does not apply the second version of the implementation. Since the electric current of the electric motor 6 can be reduced, it is possible to further reduce the losses of the inverter 2 and the motor 6.

Note options exercise of the purpose of the explanation is a device for converting power for excitation of the motor, which controls a synchronous motor with permanent magnets. However, the control method explained above can be applied to devices that convert power for excitation of the motor, which is controlled so as to energize the motors of other types.

The configuration explained in the variants of implementation, are u what Karami content of the present invention. It goes without saying that the configuration can be combined with other well-known technologies, and are subject to change, for example, through the elimination of parts without departure from the essence of the present invention.

Additionally, this detailed description mainly explains the application to the device power conversion for the excitation of the electric motor for a vehicle with an electric motor. However, the scope is not limited to this. Needless to say, that may also apply to other areas of industrial applications.

Industrial applicability

As explained above, the device power conversion for the excitation of the motor according to the present invention is useful as an invention that permits the management of suppression for 2f-component of the power source while suppressing the occurrence of a current overload and excessive ripple torque in the AC motor.

Explanation of reference designations

1 - capacitor

2 - second unit of power conversion (inverter)

3, 4, 5 - sensor current

6 - motor

7 - rotation sensor

8 - sensor voltage

10 - forming unit control commands current

11 - forming unit control commands underlying the current d-axis

14 - adder

15 - shaping unit command current control q-axis

20 is a control block current d-axis

21 - calculation module isolation of the q-axis

the 22 - unit calculation of noise immunity d-axis

23 is a control block current q-axis

30 - unit calculation of the coefficients of the modulation

40 - unit calculation of governors of the phase angles

50 - forming unit control commands voltage/signals pulse-width modulation

55 is a block computation command voltage

57 - forming unit asynchronous multi-pulse carrier signal

58 - forming unit synchronous rahimullah bearing

59 - selection switch

60 - processor switch pulse mode

61-63 comparator

64-66 - inverting schema

69 - computing unit angular frequency inverter

70 - multiplier

71 - unit calculating a signal pulse suppression

72 - bandpass filter (BPF)

73 - adder

74 - divider

80 - unit control commands control the current

81 - unit limit

82 - amplifier

84 - unit subtraction

85 - forming unit control commands modulation factor of

the 90 - unit conversion three-phase coordinates in the dq-axis

95 - unit calculating a reference phase angles

100 - second control unit

150 - unit calculating indexes voltage amplitude

200 - per the first control unit

210 - forming unit control commands DC voltage

211 control unit voltage

212 - unit current control

213 - block signal pulse-width modulation

214 - current sensor

220 - the first conversion unit capacity (inverter)

230 - source AC power

240 - table of control commands DC voltage

250 - table of control commands modulation factor of

251 - block subtraction

252 - unit limit

253 - unit proportional integration

254 - adder

280 - unit control DC voltage

300 - conversion device power for excitation of the motor.

1. The device power conversion for the excitation of the motor, comprising:
the first block (220) power conversion, which is connected to the source (230) AC power and converts the AC voltage from the power source AC into a DC voltage;
the second block (2) power conversion, which is connected to the first block (220) power conversion and converts the DC voltage into AC voltage and outputs the AC voltage to the electric motor (6) AC;
the first unit (200) controls, which manipulated the first block (220) power conversion; and
the second control unit (100)that controls the second block (2) power conversion, thus:
the second unit (100) includes:
block (10) teams ' formation control current, which is generated based on at least the control command torque command current control for AC motor;
block (150) calculating indexes voltage amplitude, which calculates on the basis of control commands by the current index of the amplitude of the voltage that must be applied to the motor (6) AC;
block (80), the control commands of the control current, which is generated based on at least the index of the voltage amplitude and frequency of the AC motor, the control value of the command current control to regulate the control command current; and
block (71) the signal suppression pulse, which is generated based on the DC voltage signal (BTPMFCMP) suppression of ripples, and
the second block (100) control forms based on the control signal includes a command current control, adjustable by value (dv) control commands of the control current signal (BTPMFCMP) pulse suppression, the signal pulse-width modulation in the second block (2) power conversion and outputs the signal latitudes is about width modulation.

2. The device according to claim 1, in which the second block (100) controls, when the frequency of the AC motor is in a predefined range, the voltage output by the second conversion unit of power, equal to a predefined value, less maximum voltage that can be displayed in accordance with the DC voltage.

3. The device according to claim 2, in which the second block (100) control selects, when the management to ensure coincidence of torque generated by the AC motor, with the command torque control performed by the unit (10) teams ' formation of the current control and block (80), the control commands control the current command of the current control on the fixed-line torque based on the control command torque and the inner side of the line tension.

4. The device according to claim 1, in which the block (80), the control commands of the control current includes a block (85) formation of task teams amplitude control voltage, which generates, based on the frequency of the AC motor, the target command amplitude control voltage indicating the maximum value of the voltage amplitude.

5. The device according to claim 4, in Katoomba (80), the control commands of the control current generates on the basis of a deviation between the target command to control the voltage amplitude and the index of the voltage amplitude, the control value of the command current control.

6. The device according to claim 4, in which the block (85) formation of task teams amplitude control voltage generates, when the frequency of the AC motor is in a predefined range, the target command amplitude control voltage to set the output voltage of the second unit (2) power conversion, equal to a predefined value, less maximum voltage that can be displayed in accordance with the DC voltage.

7. The device according to claim 1, in which:
block (10) teams ' formation control current generates a first control command current d-axis, which is the current component of the magnetic flux of the AC motor, torque control, adjusts the first control command current d-axis according to the value of the control command of the control current to generate the second control command current d-axis, and generates, based on the command torque control and the second control command current d-axis first command current control q-axis, which is the current component of the torque, and
block (150) calculating indexes voltage amplitude calculates, on the basis of the second control command current d-axis and the first team pack is Alenia current q-axis the index of the voltage amplitude.

8. The device according to claim 1, in which:
the second unit (100) includes:
unit (60) of the switching pulse mode, which switches the pulse mode of the second block (2) power conversion; and
block (59) select the pulse mode, which selects according to the control unit (60) of the switching pulse mode, at least one of the multiple pulse modes, including asynchronous pulse mode to generate a signal pulse-width modulation asynchronous with the frequency of the AC motor and the synchronous pulse mode to generate a signal pulse-width modulation synchronous with the frequency of the AC motor, and
the second control unit selects, when the frequency of the AC motor is in a predefined range, centered around a frequency twice the frequency of the power source of alternating current, a synchronous pulse mode as pulse mode.

9. The device according to claim 1, in which:
the second unit (100) includes:
unit (60) of the switching pulse mode, which switches the pulse mode of the second block (2) power conversion; and
block (59) select the pulse mode, which selects according to the management block transfer is your pulse mode, at least one of the multiple pulse modes, including asynchronous pulse mode to generate a signal pulse-width modulation asynchronous with the frequency of AC induction motors and synchronous trichinosis mode for the signal pulse-width modulation, the number of pulses whose voltage half-period equal to three, formed synchronously with the frequency of the AC motor, and
the second block (100) control selects, when the frequency of the AC motor is in a predefined range, centered around a frequency twice the frequency of the power source of alternating current, synchronous trichinosis mode as pulse mode.

10. The device according to claim 8, in which the block (59) select the pulse mode selects on the basis of at least the index of the amplitude of the voltage, which does not include the signal suppression pulse, pulse mode.

11. The device according to claim 9, in which the block (59) select the pulse mode selects on the basis of at least the index of the amplitude of the voltage, which does not include the signal suppression pulse, pulse mode.

12. The device according to claim 2, in which the predetermined range is a range centered around a frequency two times higher than the second frequency source of AC power.

13. The device according to claim 2, in which a predetermined value is a value equal to or greater than 90% and lower than 100% of the maximum voltage that can be output at a constant voltage by the output voltage of the second unit (2) power conversion.

14. The device power conversion for the excitation of the motor, comprising:
the first block (220) power conversion, which is connected to the source (230) AC power and converts the AC voltage from the source (6) AC power into a DC voltage;
the second block (2) power conversion, which is connected to the first block (220) power conversion and converts the DC voltage into AC voltage, and outputs the AC voltage to the AC motor;
the first unit (200) that controls the first power conversion power; and
the second unit (100) that controls the second block (2) power conversion, thus:
the second unit (100) includes:
block (10) teams ' formation control current, which is generated based on at least the control command torque command current control for the motor (6), AC; and
block (150) calculations in which exof voltage amplitude, which calculates on the basis of control commands by the current index of the amplitude of the voltage that must be applied to the AC motor, and
the first unit (200) includes:
block (210) teams ' formation control DC voltage, which generates a command to control the DC voltage, which is the target value of the DC voltage; and
block (280) control voltage, which generates the control signal so that the DC voltage and the control voltage coincide with each other, and
management team the DC voltage, when the frequency of the AC motor is in a predefined range, is formed greater than management command DC voltage generated when the frequency of the AC motor is not in the predefined range.

15. The device according to 14, in which the block (210) teams ' formation control DC voltage generates based on the frequency of the AC motor command control DC voltage.

16. The device according to 14, in which the block (210) teams ' formation control DC voltage forms-based index of the amplitude on the order management team the DC voltage.

17. The device according to 14, in which the block (210) teams ' formation control DC voltage generates based on the frequency of the AC motor and the index of the amplitude of the voltage command control DC voltage.

18. The device according to 14, in which the block (210) teams ' formation control DC voltage generates based on the frequency of the AC motor current target value indicating the maximum index of the amplitude of the voltage, and generates on the basis of the target value that specifies the maximum index of the voltage amplitude and index of the amplitude of the voltage command control DC voltage.

19. The device 14, where the predetermined range is a range centered around a frequency twice the frequency of the AC current.

20. The device according to 14, in which a predetermined value is a value equal to or greater than 90% and lower than 100% of the maximum voltage that can be output at a constant voltage by the output voltage of the second unit (2) power conversion.



 

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3 dwg

FIELD: electricity.

SUBSTANCE: in frequency converter control method, when supply voltage fails, motor (7) rotates at the runout. When the network (1) is being restored, shaper (11) supplies signals to two (or three) channels (transistors) in various groups of inverter (6); current that exceeds the set point (approximately 200-300% of nominal motor current (7)) rises in two phases of motor. Under action of stator current pulse in rotor of motor there created is decaying current pulse and magnetic field that corresponds to it, which induces e.m.f. in stator windings, the frequency of which is equal to rotation frequency. Frequency of that signal is measured with unit (9) and transmitted to inverter (6) control unit (13). Repeated acceleration of electric motor (7) starts exactly from that frequency.

EFFECT: simplifying the construction and improving reliability.

2 dwg

FIELD: electricity.

SUBSTANCE: single-phase network-driven frequency speed regulator for a three-phase asynchronous short-circuited electric motor belongs to devices for launch and regulation of speed of three-phase asynchronous electric motors in case of power supply from single-phase network and is intended for usage in an electric drive for control of speed of three-phase asynchronous electric motors stator windings whereof are delta-connected. Each of the three reverse semiconductor commutators of the device contains two oppositely connected transistors intended for power supply of the electric motor stator windings in case of the windings delta-connection.

EFFECT: provision for the possibility to perform regulation of the electric motor rotation speed both over and below the rated speed, mean torque increase and energy performance improvement.

6 dwg

FIELD: electricity.

SUBSTANCE: linear asynchronous electric drive 1 contains an inductor consisting of a core including a heelpiece 2 and teeth 3. The heelpiece 2 is designed in the shape of a rectangular lattice at the points whereof the teeth 3 are positioned. The electrically conductive secondary element 4 is positioned on ball rests 5. The inductor winding 6 is designed to be single-phase. Each teeth has three loops positioned in its butt-end part, all the loops outlets connected to circuit closer devices such as reed relays. The first pair of loops 7 are positioned parallel to each other while the other pair of loops 8 is perpendicular to the first one. The pairs of loops 7 and 8 are intended for shielding the tooth edge parts. The coils 9 of the reed relays are connected to the commutation device 10 controlling the magnetocontrollable contacts 11 of the reeds. The single-phase winding 6 coils are connected to a single-phase voltage source via the master switch 12. Fixation of the electrically conductive secondary element 4 may be performed by four longitudinal or four cross-wise rows of shielded teeth 3 with the loops shielding the last tooth in the row opened to provide for the secondary element pitch.

EFFECT: provision for pitched relocation in the longitudinal and the cross-wise direction during power supply to the inductor winding from the single-phase voltage source.

8 dwg

FIELD: electricity.

SUBSTANCE: method of immediate response control of voltage Uc or current Ii at the output of a strip filter RLC includes operations, where: the specified current value Iuc is calculated (92) at the average force Iu of DC current Iu, flowing through the first output point of the filter between moments ti and ti+1, at the same time the specified set value Iuc is established based on equations of filter discrete conditions so that the voltage Uc or linear current I1 are equal to the preset specified value of voltage U1c or linear current Iic at the moment ti+1; an electric converter is controlled (100) to produce a current Iu, which passes through a filter, and the average force Iu of which between the moments ti and ti+1 is equal to the set value of current Iuc.

EFFECT: provision of a quick response to sharp variations in linear voltage or an opposing torque.

19 cl, 8 dwg

FIELD: electricity.

SUBSTANCE: induction motor is controlled by control of an output voltage of an inverter converting DC into AC with frequency tuning at the controlled voltage by variation of a modulation depth, which is carried out by a command to vary a magnetising component of current in a stator winding of the induction motor, where voltage from the inverter is supplied, and by a command to vary a torque-shaping component, including measurement of phase currents and an angular position of a rotor shaft within a time period equal to duration of a pulse transision function of rotor current attenuation, values of rotor current vector components relative to a stator current vector are calculated using formulas given in application materials. The device comprises an active current setter, a magnetic current setter, a vector comparison unit, a field module controller, a phase angle controller and an angle summator, an inverter control circuit, an inverter, current metres, an induction motor, an angular position metre, a vectorisation unit, a metre of a module and an angle of a stator current, the first and second units of vector rotation, the vector dynamic link, a unit to subtract angles and a unit of angle conversion.

EFFECT: increased accuracy of induction motor torque control.

4 dwg

FIELD: electricity.

SUBSTANCE: in a single-key electric drive a DC voltage source consists of two independent voltage sources (1) and (2), at the same time the plus of a source (1) is connected to the plus of a capacitor (3), with the start of a part (4) of a winding and a cathode of a diode (5). The minus of the source (1) is connected with the first output of a semiconductor key (6), with the minus of the capacitor (3) and with the start of a part (7) of the winding. The plus of a source (2) is connected to the second output of the semiconductor key (6), with the plus of a capacitor (8) and the end of the part (4) of the winding. The minus of the source (2) is connected with the minus of the capacitor (8), the end of the winding part (7) and the diode (5) anode.

EFFECT: higher reliability of an electric drive by elimination of asymmetry of supply to parts of a stator winding in a valve-inductor motor.

2 cl, 2 dwg

FIELD: transport.

SUBSTANCE: invention relates to control over AC motor electrical and mechanical characteristics. Proposed method consists in setting two new motor operating modes. First mode includes that of motor maximum torque. Second mode consists in maximum electric power saving. Said modes are loaded into parameter comparator. Number of idle work phases is defined to generate signal of idle phase set. Signals are allowed for in generating sets of the first mode work phase at maximum motor torque and second maximum power saving mode. Proposed device comprises power saving mode work phase ACS unit, idle phase quantity estimation unit, unit to compute idle phase set, unit for separate transmission of motor maximum torque signal and that of electric power maximum saving.

EFFECT: increased motor shaft torque and generated signal quality.

2 cl, 3 dwg

FIELD: electricity.

SUBSTANCE: induction motor is controlled by control of an output voltage of an inverter converting DC into AC with frequency tuning at the controlled voltage by variation of a modulation depth, which is carried out by a command to vary a magnetising component of current in a stator winding of the induction motor, where voltage from the inverter is supplied, and by a command to vary a torque-shaping component, including measurement of phase currents and an angular position of a rotor shaft within a time period equal to duration of a pulse transision function of rotor current attenuation, values of rotor current vector components relative to a stator current vector are calculated using formulas given in application materials. The device comprises an active current setter, a magnetic current setter, a vector comparison unit, a field module controller, a phase angle controller and an angle summator, an inverter control circuit, an inverter, current metres, an induction motor, an angular position metre, a vectorisation unit, a metre of a module and an angle of a stator current, the first and second units of vector rotation, the vector dynamic link, a unit to subtract angles and a unit of angle conversion.

EFFECT: increased accuracy of induction motor torque control.

4 dwg

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