Mode and an arrangement for definition of logarithmical likelihood ratio with preliminary coding

FIELD: the invention refers to the field of radio liaison particularly to arrangements and modes for definition of logarithmic likelihood ratio for turbo codes and the metrics of embranchment for convolutional codes at using preliminary coding.

SUBSTANCE: the technical result is reduction of the effect of multiplication of errors achieved because multitude of signal elements is received. At that the signal element contains the totality of modulation symbols out of the totality of coded bits, the first subset of signal elements for which the bit has the first meaning and the second subset of signal elements for which the bit has the second meaning are defined. At that the first and the second subsets are signal elements out of an extended signal group. The probability that the bit equals the first meaning or the second meaning depending from the received signal element is defined, then the symbol of flexible decision of possibility that the bit equals the first meaning or the second meaning is defined. At that the symbols of flexible decision may be represented by the logarithmical likelihood ratios.

EFFECT: reduces effect of multiplication of errors.

38 cl, 7 dwg

 

This application claims the priority of provisional patent application U.S. No. 60/334363 of 29 November 2001 on "Turbomotive with advanced coding for multipath propagation channels with fading", incorporated into the present application by reference.

The technical FIELD

The invention in General relates to radio communications. In particular, the invention relates to devices and methods for determining the logarithmic likelihood ratio (LLR) for turbocodes and metrics branching for convolutional codes using pre-encoding.

PRIOR art

Communication systems are widely used to provide different types of information exchange, such as for telephony, packet data, etc., These systems can be based on multiple access, code division multiple access (CDMA), multiple access with time division multiplexing (TDMA), orthogonal seal with frequency division (OFDM) or some other multiple access methods.

High speed data transfer with high-performance spectrum radio with strong fading due to multipath propagation is a complex task. Currently, the effective modulation method for such a channel is OFDM. The OFDM method adopted in a number of standard is tov several local area networks (LAN). The OFDM method often offer systems for broadband wireless access (BWA). Although OFDM modulation is really quite effective when transmitting over the air with strong fading due to multipath propagation, this modulation method has several shortcomings.

The drawback of OFDM systems is a utility load associated with the use of protective tones in the frequency domain cyclic prefix in the time domain. Low efficiency is also due to a permissions issue block of transmitted data. The minimum block length for transmission is equal to the number of bits per OFDM symbol. This number may be large if the number of bearing great and uses the alphabet modulation of a high order. Because the system packet data length of the frame, as a rule, is not celebratey the number of bits per OFDM symbol, the bits wasted spent on filling. The filling can significantly reduce efficiency, especially for a small length of the frame.

Another significant disadvantage of OFDM is the high sensitivity to nonlinear distortion and phase noise. The amplitude of the OFDM signal is characterized by a Gaussian distribution. The high ratio of the peak to average power of the OFDM signal makes the signal is sensitive to nonlinear distortions and unilateral constraint, because the peaks of whitefish is Ala can sometimes get in the saturation region of the power amplifier. This increases the ratio of bit errors (BER) and adjacent channel interference. Therefore, in order to weaken the effect of reducing the quality of the signal you want to go to a higher power loss.

Application of the method is OFDM with effective codes of the channel partially alleviates some of the problems described above. Channel coding in combination with a channel block interleave also eliminates the need for the OFDM system in the bit loading. However, channel coding does not solve the problem of efficiency of the OFDM system. Improper choice of OFDM parameters can lead to a noticeable decrease in the efficiency of data transfer.

The transmission system on a single carrier with limited bandwidth and quadrature amplitude modulation (QAM) high order is widely used for high-speed data transmission with high spectral efficiency over wired links and wireless systems operating within line of sight. This method does not have the aforementioned drawbacks of the OFDM method. However, in a system with a single carrier and a channel with strong fading due to multipath propagation is difficult correction channel. Linear corrector does not provide satisfactory performance. The simulation showed that even if the transmission system on the same carrier used low-speed code Cana is a, to make the total service load or spectral efficiency is the same in the transmission system on a single carrier and OFDM system, the characteristics of the system with transmission on one carrier connecting linear corrector and the ideal offset only slightly higher than OFDM.

It is known that concealer with decision feedback (DFE) is a very effective way of correcting channel problems such as severe intersymbol interference (ISI). The application of the DFE requires estimates of the last transmitted symbols without delay to subtract the value of the ISI introduced their current characters. If the last transmitted character is free from errors, the value of the brought ISI can be deducted in full, without amplification of noise. This explains the excellent performance ideal DFE, i.e. system, which implies the possibility of receiving in the receiver reliable estimates of the last transmitted symbols. If the last character made the wrong decision, it is possible reproduction errors. The simulation showed that in a channel with significant multipath propagation occurs so strong the propagation of errors that the characteristics of the DFE is below the linear characteristics of the offset.

Proposed several ways to mitigate the propagation of errors in the DFE. According to some the way I propose to give a criterion of the reliability of each adjusted flexible character. Evaluation of symbol-transferable back to the DFE based on this reliability. For example, if the adjusted symbol is characterized by high reliability, feedback is passed a tough decision, but otherwise, feedback is passed to the adjusted symbol without tough decisions.

Another way, in accordance with which I propose to iterate in the circuit between the corrector and the channel decoder mode, similar to the turbo mode, called in the literature "turboturtle". The method is essentially based on the fact that, if the channel decoder produces more reliable estimates of the coded bits at the output compared to what it gets from the offset at the input, the evaluation can be fed back to the DFE. As a result, at run time in the DFE next iteration, the degree of propagation of errors in the DFE will decrease and so the First way does not lead to a technical complication in the implementation, while the second method is much more complex implementation and slower. Unfortunately, it was found that these methods are poorly help in solving the problem of propagation of errors.

Therefore, in this technical field, there is a need in the weakening effect of the propagation of errors.

The INVENTION

The present invention relates to a device and method of recovery Yes the data transmitted in the communication system, which weaken the effect of the propagation of errors. Accepted set of symbols modulated signal (hereinafter, the modulation symbols) from the set of coded bits. The claimed device and method for recovering data transmitted in the communication system. Taken a lot of signal elements, while the signal element contains the set of modulation symbols from the set of coded bits. Define the first subset of signal elements, for which the bit has a first value and a second subset of signal elements, for which the bit has the second value. The first and second subsets represented by signal elements from the extended signal group. In accordance with the embodiment, the extended signal group expand by adding 2Mito each element in the source group, where M represents the number of signal elements in the basic one-dimensional signal group, and i is an integer.

Is determined by the probability that a bit is a first value or a second value depending on a received signal element. You can then define symbol flexible solutions for the probability that the bit is the first value or the second value. Characters flexible solutions can be represented by a log is imicheskij relations likelihood.

When channel coding, which uses a flexible solution to compute the bit LLR for turbocodes (or bit metric branching to decode algorithm Viterbi convolutional codes when making flexible decisions), coagulation adopted by the group (modulo function) before the calculation of the bit LLR or metric branching leads to a significant deterioration of the characteristics of the decoder. Essentially, LLR determined using the advanced signal group, which significantly improves the quality of the decoder.

Below is a detailed description of the various aspects and embodiments of the present invention. The following detailed description of the present invention additionally contains information about the methods, ways, receivers, transmitters, and other devices and elements that implement various aspects, embodiments of and features of the invention.

BRIEF DESCRIPTION of DRAWINGS

Signs, the nature and advantages of the present invention are explained in the detailed description given below with reference to the drawings, in which to designate the corresponding elements used identical reference position and where

figure 1 shows a simplified block diagram of a communication system, which can be implemented in various aspects and embodiments of of the Britania;

on figa and 2B shows a block diagram of the two transmitting devices that encode and modulate the data, respectively, in (1) one scheme of coding and modulation, and (2) separate coding schemes and modulation for each antenna;

figure 3 shows a block diagram of a communication system containing a pre-coder;

figure 4 shows a block diagram of a communication system that uses turbomotive and pre-encoding;

figure 5 shows an example of the accepted group of signals on the module and the extended signal group; and

figure 6 shows a diagram of the sequence of operations for determining symbol flexible solutions.

DETAILED DESCRIPTION of PREFERRED embodiments of the INVENTION

Pre-coding is a widely known way to eliminate the effect of the propagation of errors and achieve the characteristics of an ideal corrector with decision feedback (DFE). The essence of pre-coding is as follows. The perfect device DFE requires perfect channel estimation, and the last transmitted character. The receiver can obtain an almost perfect estimate of the channel, but is not able to provide an assessment of the last transmitted symbols. On the other hand, the transmitter has perfect information on the last transmitted character. Therefore, if peredach what it could obtain an estimate of the channel, then it would be possible to pre-correction channel. In terms of local radio networks (WLAN) or regional radio network (WAN), in which the station access and subscriber are in fact stationary or slowly moving, the radio channel can be considered bi-directional. At the same time as station access and subscriber feature estimates the channel because the channel is identical in both directions. If the assumption of bidirectional nature of the channel becomes invalid for some reason, the pre-coding still remains a useful solution. Channel estimation can measure and transmit back to the transmitter from the receiver during the initial session connection before transmitting data. The disadvantage of pre-correction problem is likely to increase the transmission power, as well as the probable increase of the peak-to-average capacity. However, this problem very efficiently solved by pre-coding algorithm of Tomlinson-Harashima (TH).

Figure 1 shows a simplified block diagram of communication system 100, which can be implemented in various aspects and embodiments of the present invention. In this embodiment, communication system 100 is a CDMA system, corresponding to the cdma2000, W-CDMA, IS-856 and/or other system standards CDMA Transmitting device 110 transmits data typically, the blocks from the source 112 to the data processor 114 transmitted data (TX data), formats, encodes and processes the data to generate at least one analog signal. After the analog signals are sent to a transmitter 116, which (quadrature method) modulates, filters, amplifies, and converts the frequency to a higher frequency signal(s) to generate a modulated signal. Then the modulated signal is transmitted, at least one antenna 118 (figure 1 shows one antenna, at least one receiving device.

In the receiving device 130, at least one antenna 132 (the figure also shows only one antenna) receives the transmitted signal and forwards it to the receiver 134. In the receiver 134 received signal is amplified, filtered, converted in frequency to lower frequency (quadrature), is demodulated and digitized to form samples. Then the samples are processed and decoded in the processor 136 of the received data (RX data) to recover the transmitted data. The receiving device 130 performs the processing and decoding using the method further way of processing and encoding of the transmitting device 110. Then the restored data is transmitted to the receiver 138 the data.

On figa shows the block diagram of the transmitting device 200a, which is an implementation option section of the transmitter is depicted in figure 1 of the transmitter system 110. In accordance with this embodiment only the encoding scheme is used for all NTtransmitting antennas, and the only modulation scheme is used for all NFfrequency subchannels of all transmitting antennas. The transmitting device 200a includes (1) processor 114a TX-data, which receives and encodes data in accordance with a particular encoding scheme for the formation of the encoded data, and (2) the modulator 116a, which modulates the coded data in accordance with a particular modulation scheme for the formation of the modulated data. In accordance with this processor 114a TX-data and modulator 116a are one of the embodiments respectively of the processor 114 TX-data and modulator 116, presented in figure 1.

In a specific embodiment, presented on figa, the processor 114a TX-data contains the encoder 212, channel block 214 alternation and the demultiplexer 216. The encoder 212 receives and encodes data traffic (i.e. data bits) in accordance with the selected coding scheme for the formation of the coded bits. Coding increases the reliability of the ish data. The selected coding scheme may include any combination of the encoding of the cyclic redundant check code (CRC-coding), convolutional coding, turbomotive, block coding, etc. the Following is a description of several variants of encoder 212.

Next channel block 214 interleaves interleave the coded bits according to a specific scheme of alternation and generates coded bits with alternation. Interleaving provides a temporary separation of coded bits, allows transmit data based on the average relationship of signal to noise and signal to noise (SNR) in the frequency and/or spatial subchannels used to transmit the data serves as a tool against fading and additionally eliminates the correlation between coded bits used to form each modulation symbol. Interleaving may also provide additional frequency diversity, if the coded bits are transmitted over multiple frequency subchannel. Below is a description of the coding and channel interleave.

Further, the demultiplexer 216 parts subjected to interleaving and the coded data on the NTstreams of encoded data for NTtransmitting antennas that should transmit the data. Next, NTstreams of encoded data direction is controlled in the modulator 116a.

In the specific embodiment shown in figa, modulator 116a contains NTthe OFDM modulators, and each of the OFDM modulators is intended for processing the stream of encoded data for a single transmitting antenna. Each OFDM modulator contains block 222 of display characters, block 224 inverse fast Fourier transform (IFFT) and generator 226 cyclic prefix. In accordance with this embodiment all NTblocks 222a-222t display characters implement the same modulation scheme.

Each OFDM modulator block 222 of display characters display the resulting coded bits to modulation symbols for (up to) NTfrequency subchannels for which data must be transmitted to the transmitting antenna relating to the OFDM modulator. A specific modulation scheme, to be implemented in block 222, the display of symbols is determined by controlling the modulation that is performed by the controller 130. When using OFDM modulation can be accomplished by grouping sets of q coded bits for the formation of non-binary symbols, and displaying each non-binary symbol to a specific element in the signal group corresponding to the selected modulation scheme (such as quadrature phase-shift keying (QPSK), a multilevel phase-shift keying (M-PSK), no is unavai quadrature amplitude modulation (M-QAM), or any other scheme). Each mapped signal element corresponds to the M-ary (base M) modulation symbol, where M=2q. Next, block 222 of display characters gives the vector consisting of (up to) NFthe modulation symbols for each period of transmission symbols, the number of modulation symbols in each vector corresponds to the number of frequency subchannels to be used for data transmission in the transmission period characters.

If the receiving system performs a normal non-iterative inverse mapping and decoding, the symbols appropriate to apply the display gray code, because this code provides a higher performance ratio of bit errors (BER). When displaying code gray neighboring elements in the signal group (horizontal and vertical) differ only on one of the q bit positions. Display code warming reduces the number of bit errors for events with a higher probability of errors, which correspond to the adopted modulation symbol displayed in a position near the right side, in this case only one coded bit will be received with error.

Next, the IFFT block 224 performs inverse fast Fourier transform of each vector of modulation symbols in the representation in the time region is t (which is called an OFDM symbol). The IFFT block 224 may be executed to implement the inverse transform with respect to any number (for example, 8, 16, 32, ..., NF, ...) of frequency subchannels. In accordance with the embodiment of the generator of the cyclic prefix 226 repeats part of the OFDM symbol of each OFDM symbol to form a corresponding transmitted symbol. When using cyclic prefix transmitted symbol retains its orthogonal properties in the presence of variation in the delay of multipath propagation, thereby improving performance in the face of such adverse impacts tract, as the variance of the channel due to frequency-selective fading. The transmitted symbols from the generator of the cyclic prefix 226 are received in the corresponding transmitter 122 and processed to generate a modulated signal, which is transmitted to the corresponding antenna 124.

On FIGU shows the block diagram of the transmitting device 200b, which is another embodiment of the transmitter section the transmitter system 110 depicted in figure 1. In accordance with this embodiment for each of the NTtransmitting antennas using a single coding scheme, and for all NFfrequency subchannels of each transmitting antenna using a different modulation scheme (i.e., the principle of RA is practical coding and modulation for each antenna). Specific coding scheme and modulation that should be explored for each transmitting antenna can be selected based on expected channel conditions (for example, the receiving system and transmitted back to transmitter system).

The transmitting device 200b includes (1) processor 114b TX-data, which receives and encodes data in accordance with a separate coding schemes for forming coded data, and (2) the modulator 116b, which modulates the coded data in accordance with individual modulation schemes for the formation of the modulated data. The processor 114b TX-data and modulator 116b are another variant implementation, respectively, of the processor 114 TX-data and modulator 116, presented in figure 1.

In a specific embodiment, presented on FIGU, the processor 114b TX-data contains a demultiplexer 210, NTcoding devices 212a-212t, and NTchannel units 214a-214t alternation (i.e. one group consisting of the coding device and the channel unit interleave, each transmitting antenna). The demultiplexer 210 divides the information data (i.e., information bits) to NTdata flow for NTtransmitting antennas to be used for data transmission. Next, each data stream is forwarded to the appropriate code is its structure.

Each encoder 212 receives and encodes the corresponding data stream based on a particular coding scheme selected for the corresponding transmitting antenna for forming coded bits. Next, the coded bits from each encoder device 212 are served in an appropriate channel block 214 alternation, which interleaves the coded bits based on a specific schema alternations to ensure diversity. Then channel blocks 214a-214t interleave transmit intermittent streams and encoded data for NTtransmitting antennas in the modulator 116b.

In a specific embodiment, presented on FIGU, modulator 116b comprises NTthe OFDM modulators, each modulator OFDM contains block 222 of display characters, the IFFT block generator 224 and 226 of the cyclic prefix. In accordance with this embodiment NTblocks 222a-222t display characters can implement different modulation schemes. Each OFDM modulator block 222 of display characters display groups of qncoded bits for the formation of Mn-ary modulation symbols, where Mncorresponds to a particular modulation scheme selected for the nth transmitting antenna (as determined by the controller 130, managing modulation) and Mn=2q. The description of the subsequent processing block of the IFFT 224 and the generator of the cyclic prefix 226 above.

Other structural embodiments of the transmitting device within the scope of the present invention. For example, the coding and modulation can separately run for each subgroup transmit antennas, each transmit channel or each group of transmission channels. Variants encoding device 212, channel blocks 214 interleave, blocks 222 of the display characters, IFFT blocks 224 and generators 226 cyclic prefix enough known to experts in the art and therefore not discussed in detail in the present description.

A more detailed description of coding schemes and modulation for MIMO systems with and without using OFDM contained in patent applications U.S. No. 09/826481 and No. 09/956449 "Method and apparatus for using state information of a channel in a wireless communications system", respectively, from March 23, 2001 and September 18, 2001; in patent application U.S. No. 09/854235 for "Method and device for processing data in a communication system with multiple inputs and multiple outputs using the information of channel status" on may 11, 2001; in patent application U.S. No. 09/776075 on the encoding Scheme for the system wireless" from February 1, 2001; and patent application U.S. No. 09/993087 on "communication System multiple access with multiple inputs and multiple outputs" dated November 6, 2001, the CE listed applications assigned to the owner of the rights to the present invention and is incorporated into this description by reference. You can also use other coding scheme and modulation, which is also not beyond the scope of the present invention.

A description of an example OFDM system is contained in the patent application U.S. No. 09/532492 on "high-performance communication system using a modulation of the number of carrier" dated March 30, 2000, assigned to the owner of the rights to the present invention and is incorporated into this description by reference. In addition, the description of the modulation method OFDM is given in the article John A.C. Bingham, "Multicarrier Modulation for Data Transmission: An Idea Whose Time Has Come", IEEE Communications Magazine, May 1990, which is incorporated into this description by reference.

To encode the data before sending it, you can use the encoders of different types. For example, the encoder may implement encoding any of the following codes, namely, (1) serial cascaded convolutional code (SCCC), (2) parallel cascaded convolutional code (PCCC), (3) simple convolutional code, (4) cascade code made up of a block code and a convolutional code, etc. Cascaded convolutional codes are also referred to as turbocode.

The above-described signal processing methods provide voice, video, packet data, and other types of information in one direction. Bidirectional communication system provides two-way transmission of Yes is different and operates in a similar manner.

Figure 3 shows the block diagram 300 of a communication system containing a pre-coder. Figure 3 akdenotes a complex modulation symbol 304 from a group of signals with quadrature amplitude modulation (QAM), where k is the temporal index. Considered a square group of QAM signals, which can be considered as a direct product of two groups of signals with amplitude and pulse modulation (PAM), with M elements in each group, namely, (- (M-1), -(M-2), ..., (M-3), -(M-1)). Complex modulation symbol 304 is an input symbol for the pre-coding device 308. The pre-coding device 312 is defined by the following expression:

Xk=ak-[Xk-1h-1+Xk-2h-2+...+Xk-Lh-L] modulo 2M, (1)

which can be rewritten separately for real and imaginary parts,

Xk=ak+2M(lk+jmk)-[Xk-1h-1+Xk-2h-2+...+Xk-Lh-L], (2)

where lkand mkare integers, for which the real and imaginary components of Xkare within +/- M, i.e. MRe[Xk], Im[Xk]M.

In accordance with the foregoing, the pre-coding device is a function of the current symbol (ak) minus the product of the previous pre-coding condition the device (X k-1and so on) on the previous impulse response of the channel (h-1and so on).

Then the output 312 is served in a unit of a combined transfer function 316. As can be seen from figure 4 (description see figure 4 below), H(z) figure 3 represents the combined transfer function 316 filter on the transmission side, multipath channel, the filter on the receiving side and related to the corrector filter with a direct link. Assuming that the combined impulse response is limited to L+1 symbols, the function H(z) is defined by the expression

H(z)=1+h-1z-1+h-2z-1+...++h-Lz-L] (3)

The output unit of the combined transfer function 316 denoted by Yk(or 320). Therefore, equation (2) gives

ak+2M(lk+jmk)=Xk+Xk-1h-1+Xk-2h-2+...+Xk-Lh-L=Yk(4)

Nkdenotes a complex made white noise with Gaussian distribution (AWGN) 324 with the spectral power density N0/2. When combined transfer function 320 mix made white noise with Gaussian distribution, we get a function of Zk(328)defined by the expression

Zk=Yk+Nk=ak+2M(lk+jmk)+Nk(5)

and

Wk=Zkmod 2M (6)

where the function MOD 2M 332 means limiting the energy transferred si is Nala closer to the energy of an unexpected signal group, and Wk(336) denotes statistics solutions.

In accordance with the foregoing, the use of pre-coding leads to an extension of the original signal group. This means that, if akrepresents a signal element in the initial group of QAM signals, ak+2M(lk+jmkalso is reliable signal element in the extended signal group, where lkand jmkare integers. Essentially, the operation constraints modulo (modulo 2M) in the receiver coils extended signal group back to the original signal group.

Characteristics of pre-coding is slightly worse than the ideal characteristics corrector type DFE, at least for the following reasons: the signal after the pre-coding is no longer discrete, but becomes uniformly distributed in the interval [-M, M], which leads to a few more of the transmitted energy at the same minimum distance between two signal elements. This effect is known as loss in the preliminary coding, the value of which is determined by the expression

These losses become negligible for large signal group. In addition, the characteristics of the pre-coding bit worse than characterized the Tiki ideal corrector type DFE, because pre-coding leads to an extension of the original signal group, the average number of nearest neighbors increases, which leads to increased error rate. However, the pre-coding is a very powerful, simple and convenient means of achieving the characteristics of an ideal DFE.

The block diagram of communication system 400 that uses turbomotive and pre-coding, shown in figure 4. The transmitted block of binary data 404 is encoded turbodiesel device 408, which generates a sequence of code bits 412. Turbo code can be parallel or serial cascade code. In addition, you can use the "piercing" (delete items) code to form any code rate. After turbomotive sequence of code bits 412 is supplied in block 416 of the display, where the bits are grouped (2log2M) and displayed in the element in the group of signals with modulation of the form M2-QAM. In the embodiment, use gray codes. The output of block 416 display is formed by a sequence of complex modulation symbols 420. A sequence of complex modulation symbols 420 is supplied in pre-encoder 424. The operation of the pre-coding device described will apply the flax to 3.

At the output of pre-coding device also forms a complex quantity 428. In accordance with the embodiment, the integrated signal 428 contains the real and imaginary components, uniformly distributed in the interval between-M and+M, where M denotes the number of signal elements in a component of the group of signals with amplitude and pulse modulation (PAM). Then the output signal 428 pre-coding device is in a formative impulses filter 432 on a transmitting side of the system. Filter 436 at the receiving side is more formative impulses filter in the receiver. Filter 432 on the transmission side and the filter 436 at the receiving side can be such filters Nyquist, using the algorithm to compute the square root, which provide a combined response of the form of Nyquist pulse.

The transmitting channel 440 local radio network can be modeled as independent Rayleigh channel fading due to multipath propagation, which brought additive white noise with Gaussian distribution (AWGN) 444. Filter 448 with a direct link is a section of a direct communication channel corrector and functionally can be fractional interval. Filter 436 at the receiving side in combination with a filter with a direct link can be considered as equivalent to comb kirovenego channel matched filter with noise whitening filter. If you know the filter's parameters on the transmit and receive sides of the system and the impulse response of the channel, the filter coefficients with a direct link and a pre-coding device can be calculated using the criterion of minimum mean square error (MMSE).

Zndenotes the output signal 452 filter with a direct link, which is served in the transmitter 456 metric LLR (n is the time index). The function evaluator 456 LLR metrics can perform a microprocessor, software, firmware, executable in a microprocessor embedded in a specialized integrated circuit (ASIC), or any other means. The output 460 of the transmitter 456 LLR metrics indicate the likelihood that a particular bit has a particular value, and served in a cascaded convolutional encoder 464, for example turbocoding device that generates decoded data 468. The filter output signal 448 with a direct link is determined by the expression

Zn=An+jBn=an+2M(ln+jmn)+N'n, (7)

where anis the corresponding transmitted symbol QAM signal, and N'nis a comprehensive reference made white noise with Gaussian distribution (AWGN). Znis obtained a flexible solution for the transmitted symbol an.

The transmitter 456 met the IKI calculates LLR LLR values at 2log 2M bits for each of the obtained flexible character of the QAM signal. Due to the symmetry of the square works group of QAM signals to display a gray code, the value of the LLR specific code bits depends on An(the real component) or Bn(imaginary component) and from the corresponding one-dimensional elements of the PAM signal. In other words, in order to compute the LLR values, we can assume that the received QAM signal consists of two independent signals PAM. Therefore, the LLR value for a given code bit bk(k is the index of the bits in the group log2M bits or the designation of the PAM signal; 0k<log2M), corresponding to the accepted signal Anassuming equiprobable modulation symbols is determined by the expression

whereanddenote the subset of signal elements for a signal with a multilevel amplitude-width modulation (signal M-PAM), which respectively bk=0 and bk=1. Previously stated that due to pre-coding, the obtained flexible solutions Anand Bnbelong to an extended group of PAM signals. Therefore, the LLR value is found by determining the probability that Anaccepted under the condition that s passed. As can be seen from the last h is STI equations (8), the calculation of the LLR values may include the influence of factors of noise σ2.

The operation Mod 2M on Anand Bnroll the accepted signal element in the source group, which is beneficial when Anand Bnyou should get a tough decision. However, if you use channel coding, which uses a flexible solution to calculate the value of the bit LLR for turbocodes (or bit metric branching to decode algorithm Viterbi convolutional codes when making flexible decisions), the collapse adopted by the group before the calculation of the bit LLR or metric branching leads to a significant deterioration of the characteristics of the decoder. An example of this is figure 5.

Figure 5 shows the received group of signals on the module and the extended signal group. Frame 504 is (unexpanded) signal group on the module that contains the elements of -3, -1, 1 and 3, which corresponds to the gray codes (bits b0and b1respectively 11, 10, 00 and 01. If, as shown, adopted element 508 (located directly behind "4") and the operation modulo 2M, then the element 508 is converted to item 512 (located directly in front of -4). In the unexpanded group assesses the probability that the element 512 is equal to 0 or 1. The probability that bit b0equal to "1" is very high, because only the near ISEE value for bits b 0equal to "1" (with a probability of 95%). However, if we consider the extended signal group, we also evaluate the probability that the element 508 is equal to 0 or 1. Because the element 508 a little closer to "11", than to "01", then the probability that bit b0equal to "1" is much lower (probability about 55%). Therefore, the use of advanced signal group before calculating the value of the LLR and without the use of operations modulo 2M provides a much more accurate determination of the probability for this bit.

Thus, when there is a preliminary encoding device, to calculate the value of the bit LLR or metric branching is applied modification with the exception of the operations module and calculating the value of the bit LLR to the extended signal group. In other words, manyandexpand by adding 2Mito each element in the original set, where i is an integer. Further, the LLR value is determined using the extended set. The range of possible values of i that need to be considered from many channel realizations, is predefined. Modeling is installed, which is usually sufficient to use many channel realizations, where i=-2, -1, 0, 1, 2, however, it is assumed that the can is about to use any value of i. In the assumption of the adequacy of the above range i the cardinal number of expanded signal setsandfour times more than the original set. As a result, significantly increases the complexity of calculation of LLR values. However, the mentioned complexity can be dramatically reduced if the calculation of LLR values or metrics to consider only those elements that are within ± M from the adoption of the element.

Figure 6 is a diagram 600 of the sequence of operations of the method to determine the value of LLR. At step 604 take many demodulated signal elements. The demodulated signal elements contain a set of encoded bits and the noise. At step 608 determines a first subset of signal elements and the second subset of signal elements. Then, at step 612 determines the probability that the bit is adopted, provided that it is accepted specific flexible solution. Adopted a flexible solution belongs to the extended signal group. Therefore, it follows from equation (8), the value of the LLR at the step 616 is defined as the logarithm of the relationship of the sum of probabilities that the adopted bit is "1" or "0".

Diversity antennas, for example, just as is the case in a system with multi-input and multi-channel is hodom (MIMO), and is effective circuit solution to improve the quality of data transmission on fading channel. The above method of pre-coding in combination with the definition of the LLR values using the extended bit groups are also useful for communication systems that use the spacing of multiple receiving antennas for receive diversity with summation or selection.

So, given the description of a new and improved method and device for determining the LLR values in combination with the pre-encoder. For specialists in the art it is obvious that information and signals may be represented using any of a variety of technologies and technical means. For example, data, instructions, commands, information, signals, bits, symbols, and elementary signals, which are referred to in the above description may be represented by voltages, currents, electromagnetic waves, magnetic fields or particles, optical fields or particles, or any combination of the listed objects.

For specialists in the art it is obvious that various used to illustrate the logical blocks, modules, circuits, and algorithm steps, the description of which is given in connection with the above description options implemented the program of the present invention, may be implemented as electronic hardware, computer software, or combinations thereof. To more clearly demonstrate the interchangeability of hardware and software, the above descriptions of the various used to illustrate components, blocks, modules, circuits, and steps are generalized at the level of the functions they perform. Whether these functions are implemented in hardware or software depends upon the particular application and design constraints imposed on the system as a whole. Technicians can perform the above functions in different ways in each specific case of application, however, these options implementation cannot be considered as going beyond the scope of the present invention.

Various used to illustrate the logical blocks, modules, and circuits, descriptions of which are given in connection with the above in the description of variants of implementation of the present invention may be implemented or performed using a generic processor, digital signal processor (DSP), a specialized integrated circuit (ASIC), programmable gate array (FPGA) or other programmable logic device, a logic gate IC with a low degree of integration is or transistor logic, discrete hardware components, or any combination of these elements that is intended to perform the above functions. Universal processor may be a microprocessor, or processor may be any conventional processor, controller, microcontroller, or state machine. The processor can also be implemented as a combination of computing devices, such as a combination of a digital signal processor (DSP) and microprocessor, the set of microprocessors, at least one microprocessor in combination with a base digital signal processor (DSP), or any other similar configuration.

The steps of the method or algorithm, the description of which is given in connection with the above in the description of variants of implementation of the present invention, can be embedded directly in hardware, in a software module, executable by a processor, or in a combination of hardware and software. A software module may reside in the storage device, random access (RAM), flash memory, permanent memory (ROM), erasable programmable ROM (EPROM) or electrically erasable programmable ROM (EEPROM), registers, hard disk, removable disk, optical disk (CD-ROM), or any other media, Izv the STN in the art. The processor and the appropriate storage media may be part of a specialized integrated circuit (ASIC). Diagram of the ASIC can be included in the subscriber unit or in any part of the infrastructure of the radio system. In another embodiment, the processor and the storage media can be an integral part of the discrete components in the user terminal.

The foregoing description of certain embodiments of the invention, is intended to give any expert in the art the ability to make or use the present invention. Specialists in the art of the obvious possibility of making various changes to the above embodiments of, with the above General principles can be applied to other variants of implementation without going beyond essence or scope of the present invention. Therefore, the present invention is not limited to the above variants of implementation, and corresponds to the widest extent set forth in the present description principles and new features.

1. Method of recovering data transmitted in the communication system, namely, that

take plenty of signal elements with the signal element contains a set modulation the output symbols from the set of coded bits;

determine a first subset of signal elements, for which the bit has the first value;

determine a second subset of signal elements, for which the bit has a second value when the first and second subsets represented by signal elements from the extended signal group;

determine the probability that a bit is a first value or a second value depending on a received signal element; and

define the symbol flexible solutions for the probability that the bit is the first value or the second value

this enhanced signaling group form by adding 2Mito each element in the source signal group, where M denotes the number of signal elements in the source signal group, a i is an integer.

2. The method according to claim 1, characterized in that the characters flexible solutions presents a logarithmic relationship likelihood (LLR).

3. The method according to claim 2, characterized in that the value of the LLR is determined by the following formula:

where bkrepresents a code bit, a k is the index of the bits in the group log2M bits or designation signal with amplitude and pulse modulation (PAM), where 0≤k<log2M; Anindicates the received signal corresponding to b ;anddenote the subset of signal elements for a signal with a multilevel amplitude-width modulation (M-PAM), which respectively bk=0 and bk=1.

4. The method according to claim 1, characterized in that the characters flexible solutions contain information that is transmitted via the channel, and external information.

5. The method according to claim 1, characterized in that the characters flexible solutions contain information for at least one spatial subchannel and at least one frequency subchannel used to transmit the combined modulation symbols.

6. The method according to claim 1, characterized in that the radio system is a system of orthogonal compaction with frequency division (OFDM).

7. The method according to claim 1, characterized in that the radio system is a system with multi-input and multi-output (MIMO).

8. The method according to claim 7, characterized in that the system performs MIMO orthogonal seal with frequency division (OFDM).

9. The method according to claim 1, characterized in that the extended signal group limit +/- M elements from a received item.

10. The method according to claim 1, characterized in that the source signal is a group of signals with amplitude and pulse modulation (PAM).

11. The method of determining in a radio system the ligature characters flexible solutions for the adopted modulation symbols, namely, that

determine a first subset of signal elements, for which the bit is a first value;

determine a second subset of signal elements, for which the bit is the second value, with the first and second subsets represented by signal elements from the extended signal group;

determine the probability that a bit is a first value or a second value depending on a received signal element; and

define the symbol flexible solutions for the probability that the bit is the first value or the second value

this enhanced signaling group form by adding 2Mito each element in the source signal group, where M denotes the number of signal elements in the source signal group, a i is an integer.

12. The method according to claim 11, characterized in that the characters flexible solutions presents a logarithmic relationship likelihood (LLR).

13. The method according to item 12, wherein the LLR value is determined by the following formula:

where bkrepresents a code bit, a k is the index of the bits in the group log2M bits or designation signal with amplitude and pulse modulation (PAM), where 0≤k<log2M; Andenotes adopted the th signal, the corresponding bk;anddenote the subset of signal elements for a signal with a multilevel amplitude-width modulation (M-PAM), which respectively bk=0 and bk=1.

14. The method according to claim 11, characterized in that the radio system is a system of orthogonal compaction with frequency division (OFDM).

15. The method according to claim 11, characterized in that the radio system is a system with multi-input and multi-output (MIMO).

16. The method according to item 15, wherein the MIMO system performs orthogonal seal with frequency division (OFDM).

17. The method according to claim 11, characterized in that the extended signal group limit +/- M elements from a received item.

18. The method according to claim 11, characterized in that the source signal is a group of signals with amplitude and pulse modulation (PAM).

19. Restore device in the communication system transmitted data containing

means for receiving the set of signal elements, while the signal element contains the set of modulation symbols from the set of coded bits;

means for determining the first subset of signal elements, for which the bit is a first value;

means DL is determining the second subset of signal elements, which bit is the second value, with the first and second subsets represented by signal elements from the extended signal group;

means for determining a probability that the bit is a first value or a second value depending on a received signal element; and

means for determining the symbol flexible solutions for the probability that the bit is the first value or the second value

this extended signal group generated by adding 2Mito each element in the source signal group, where M denotes the number of signal elements in the source signal group, a i is an integer.

20. The device according to claim 19, characterized in that the characters flexible solutions presents a logarithmic relationship likelihood (LLR).

21. The device according to claim 20, characterized in that the value of the LLR is determined by the following formula:

where bkrepresents a code bit, a k is the index of the bits in the group log2M bits or designation signal with amplitude and pulse modulation (PAM), where 0≤k<log2M; Anindicates the received signal corresponding to bk;anddenote the subset of signal elements for si is Nala with multilevel amplitude-width modulation (M-PAM), for which, respectively, bk=0 and bk=1.

22. The device according to claim 19, characterized in that the characters flexible solutions contain information that is transmitted via the channel, and external information.

23. The device according to claim 19, characterized in that the characters flexible solutions contain information for at least one spatial subchannel and at least one frequency subchannel used to transmit the combined modulation symbols.

24. The device according to claim 19, characterized in that the radio system is a system of orthogonal compaction with frequency division (OFDM).

25. The device according to claim 19, characterized in that the radio system is a system with multi-input and multi-output (MIMO).

26. The device according A.25, characterized in that the system performs MIMO orthogonal seal with frequency division (OFDM).

27. The device according to claim 19, characterized in that the extended signal group is limited to +/- M elements from a received item.

28. The device according to claim 19, characterized in that the source signal is a group of signals with amplitude and pulse modulation (PAM).

29. Restore device in the communication system transmitted data containing

the receiver is made with the possibility of taking multiple modulation symbols of the scoop is Prosti coded bits;

the processor associated with the receiver and configured to perform the following stages of the method:

defining a first subset of signal elements, for which the bit is a first value;

determining a second subset of signal elements, for which the bit is the second value, with the first and second subsets represented by signal elements from the extended signal group;

determining the probability that the bit is a first value or a second value depending on a received signal element; and

the symbol definition flexible solutions for the probability that the bit is the first value or the second value

this extended signal group generated by adding 2Mito each element in the source signal group, where M denotes the number of signal elements in the source signal group, a i is an integer.

30. The device according to clause 29, characterized in that the characters flexible solutions presents a logarithmic relationship likelihood (LLR).

31. The device according to clause 29, wherein the LLR value is determined by the following formula:

where bkrepresents a code bit, a k is the index of the bits in the group log2M bits or what oznaczenie signal with amplitude and pulse modulation (PAM), where 0≤k<log2M; Anindicates the received signal corresponding to bk;anddenote the subset of signal elements for a signal with a multilevel amplitude-width modulation (M-PAM), which respectively bk=0 and bk=1.

32. The device according to clause 29, wherein the enhanced signal group is limited to +/- M elements from a received item.

33. The device according to clause 29, wherein the source signal is a group of signals with amplitude and pulse modulation (PAM).

34. The computer-readable storage medium containing a sequence of machine-readable control commands by a computer system, the execution of which by a computer system performs a method of recovering data transmitted in the communication system, comprising stages

taking multiple signal elements with the signal element contains the set of modulation symbols from the set of coded bits;

determining the first subset of signal elements, for which the bit has the first value;

determine a second subset of signal elements, for which the bit has a second value when the first and second subsets represented by signal elements of EXT is adopted for signal group;

determine the probability that the bit is a first value or a second value depending on a received signal element; and

the definition of a symbol flexible solutions for the probability that the bit is the first value or the second value

this extended signal group generated by adding 2Mito each element in the source signal group, where M denotes the number of signal elements in the source signal group, a i is an integer.

35. The data carrier according to 34, characterized in that the characters flexible solutions presents a logarithmic relationship likelihood (LLR).

36. The data carrier according to clause 34, wherein the LLR value is determined by the following formula:

where bkrepresents a code bit, a k is the index of the bits in the group log2M bits or designation signal with amplitude and pulse modulation (PAM), where 0≤k<log2M; Anindicates the received signal corresponding to bk;anddenote the subset of signal elements for a signal with a multilevel amplitude-width modulation (M-PAM), which respectively bk=0 and bk=1.

37. The data carrier according to clause 34, wherein R is spider veins signal group is limited to +/- M elements from a received item.

38. The data carrier according to clause 34, wherein the source signal is a group of signals with amplitude and pulse modulation (PAM).



 

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